Method for estimating cfo in wireless lan system that uses 16 qam

ABSTRACT

Disclosed is a method for estimating a CFO, the method comprising: measuring a size ratio of a signal pair received in two consecutive OFDM symbols of a predetermined subcarrier; determining the received signal pair as a signal pair to be used for estimating a residual CFO when the size ratio is within a first threshold range; and estimating the residual CFO, using only the received signal pair which has been determined to be used for estimating the residual CFO.

TECHNICAL FIELD

Following description relates to a wireless communication system, andmore particularly, to a method of estimating a CFO in a wireless LANsystem using 16 QAM and an apparatus therefor.

BACKGROUND ART

Recently, with development of information communication technology,various wireless communication technologies have been developed. Amongothers, a wireless local area network (WLAN) enables wireless access tothe Internet using a portable terminal such as a personal digitalassistant (PDA), a laptop, a portable multimedia player (PMP) in a home,an enterprise or a specific service provision area based on radiofrequency technology.

In order to overcome limitations in communication rate which have beenpointed out as weakness of a WLAN, in recent technical standards, asystem for increasing network speed and reliability and extendingwireless network distance has been introduced. For example, in IEEE802.11n, multiple input and multiple output (MIMO) technology usingmultiple antennas in a transmitter and a receiver has been introduced inorder to support high throughput (HT) with a maximum data rate of 540Mbps or more, to minimize transmission errors, and to optimize datarate.

As next-generation communication technology, machine-to-machine (M2M)communication technology has been discussed. Even in an IEEE 802.11 WLANsystem, technical standards supporting M2M communication have beendeveloped as IEEE 802.11ah. In M2M communication, a scenario in which asmall amount of data is communicated at a low rate may be considered inan environment in which many apparatuses are present.

Communication in a WLAN system is performed in a medium shared betweenall apparatuses. As in M2M communication, if the number of apparatusesis increased, in order to reduce unnecessary power consumption andinterference, a channel access mechanism needs to be more efficientlyimproved.

DISCLOSURE OF THE INVENTION Technical Tasks

The present invention is devised to solve the above-mentioned problemsand an object of the present invention is to enable a reception moduleto estimate a CFO accurately.

Another object of the present invention is to increase accuracy ofestimating a CFO in a manner that a reception module sufficientlysecures the number of samples to be utilized for estimating the CFO.

The other object of the present invention is to apply a CFO estimationmethod of a blind type to a high order modulation scheme.

The technical problems solved by the present invention are not limitedto the above technical problems and other technical problems which arenot described herein will become apparent to those skilled in the artfrom the following description.

Technical Solution

To achieve these and other advantages and in accordance with the purposeof the present invention, as embodied and broadly described, accordingto one embodiment, a method of estimating a CFO (carrier frequencyoffset), which is estimated by a reception module in a wirelesscommunication system using 16-QAM (quadrature amplitude modulation),includes the steps of measuring a size ratio between a pair of receivedsignals received on two contiguous (or consecutive) OFDM (orthogonalfrequency division multiplexing) symbols for a specific subcarrier, ifthe size ratio belongs to a first threshold range, determining the pairof received signals as a pair to be used for estimating a residual CFO,and estimating the residual CFO using only pairs of received signalswhich are determined to be used for estimating the residual CFO.

The method can further include the step of measuring a phase differencebetween the pair of received signals. In this case, if the size ratiobelongs to the first threshold range and the phase difference belongs toa second threshold range, the determining step can determine the pair ofreceived signals as a pair to be used for estimating the residual CFO.

If the phase difference does not belong to the second threshold rangeand belongs to a third threshold range, the method can further includethe step of changing a phase of multiplication of the pair of receivedsignals as much as a predetermined value.

If the phase difference does not belong to the second threshold rangeand does not belong to the third threshold range, the method can furtherinclude the step of determining not to use the pair of received signalsfor estimating the residual CFO.

The second threshold range may indicate a range within a predeterminedvalue from one value among 0°, 90°, 180°, and 270°.

The third threshold range may indicate a range within a predeterminedvalue from one value among 46°, 136°, 226°, and 316°.

The first threshold range may indicate a range within a predeterminedvalue from one value among 1, ⅓, and 3.

The step of estimating the residual CFO can estimate the residual CFOaccording to a blind scheme which is applied when QPSK (quadrature phaseshift keying) is used.

To further achieve these and other advantages and in accordance with thepurpose of the present invention, according to a different embodiment, areception module estimating a CFO in a wireless communication systemusing 16-QAM (quadrature amplitude modulation) includes a transmitter, areceiver, and a processor configured to operate in a manner of beingconnected with the transmitter and the receiver, the processorconfigured to measure a size ratio between a pair of received signalsreceived on two contiguous OFDM symbols for a specific subcarrier, theprocessor, if the size ratio belongs to a first threshold range,configured to determine the pair of received signals as a pair to beused for estimating a residual CFO, the processor configured to estimatethe residual CFO using only pairs of received signals which aredetermined to be used for estimating the residual CFO.

Advantageous Effects

According to the embodiments of the present invention, it is able tohave the following effects.

First of all, it is able to increase a decoding efficiency of areception signal in a manner that a reception module precisely measuresa residual CFO.

Second, if the number of samples utilized for estimating a CFO isincreased, it is able to more efficiently perform a blind CFO estimationscheme.

Third, it is able to estimate a CFO by utilizing a blind type even in 16QAM corresponding to a high order compared to BPSK, QPSK, and the like.

The effects of the present invention are not limited to theabove-described effects and other effects which are not described hereinmay be derived by those skilled in the art from the followingdescription of the embodiments of the present invention. That is,effects which are not intended by the present invention may be derivedby those skilled in the art from the embodiments of the presentinvention.

DESCRIPTION OF DRAWINGS

The accompanying drawings, which are included to provide a furtherunderstanding of the invention, illustrate embodiments of the inventionand together with the description serve to explain the principle of theinvention. The technical features of the present invention are notlimited to specific drawings and the features shown in the drawings arecombined to construct a new embodiment. Reference numerals of thedrawings mean structural elements.

FIG. 1 is a diagram showing an exemplary structure of an IEEE 802.11system to which the present invention is applicable;

FIG. 2 is a diagram showing another exemplary structure of an IEEE802.11 system to which the present invention is applicable;

FIG. 3 is a diagram showing another exemplary structure of an IEEE802.11 system to which the present invention is applicable;

FIG. 4 is a diagram showing an exemplary structure of a WLAN system;

FIG. 5 is a diagram illustrating a link setup process in a WLAN system;

FIG. 6 is a diagram illustrating a backoff process;

FIG. 7 is a diagram illustrating a hidden node and an exposed node;

FIG. 8 is a diagram illustrating request to send (RTS) and clear to send(CTS);

FIG. 9 is a diagram illustrating power management operation;

FIGS. 10 to 12 are diagrams illustrating operation of a station (STA)which receives a traffic indication map (TIM);

FIG. 13 is a diagram illustrating a group based association identifier(AID);

FIGS. 14 to 16 are diagrams showing examples of operation of an STA if agroup channel access interval is set;

FIGS. 17 and 19 are diagrams illustrating frame structures according tothe present invention and constellations thereof;

FIG. 20 is a diagram illustrating frequency-domain pilot signalsaccording to the present invention;

FIGS. 21 and 22 are diagrams for explaining a method of estimating a CFOaccording to the present invention;

FIG. 23 is a flowchart for a method of estimating a CFO according to thepresent invention;

FIGS. 24 to 26 are diagrams for explaining a method of estimating a CFOaccording to the present invention;

FIG. 27 is a flowchart for a method of estimating a CFO according to thepresent invention;

FIG. 28 is a diagram illustrating a resource block according to thepresent invention;

FIG. 29 is a flowchart for a method of estimating a CFO according to thepresent invention;

FIGS. 30 and 31 are diagrams for a method of segmenting 16-QAMconstellation according to embodiments proposed in the presentinvention;

FIG. 32 is a flowchart for a method of estimating a CFO according to thepresent invention;

FIG. 33 is a diagram for configurations of a user equipment and a basestation according to one embodiment.

BEST MODE Mode for Invention

Although the terms used in the present invention are selected fromgenerally known and used terms, terms used herein may be varieddepending on operator's intention or customs in the art, appearance ofnew technology, or the like. In addition, some of the terms mentioned inthe description of the present invention have been selected by theapplicant at his or her discretion, the detailed meanings of which aredescribed in relevant parts of the description herein. Furthermore, itis required that the present invention is understood, not simply by theactual terms used but by the meanings of each term lying within.

The following embodiments are proposed by combining constituentcomponents and characteristics of the present invention according to apredetermined format. The individual constituent components orcharacteristics should be considered optional factors on the conditionthat there is no additional remark. If required, the individualconstituent components or characteristics may not be combined with othercomponents or characteristics. In addition, some constituent componentsand/or characteristics may be combined to implement the embodiments ofthe present invention. The order of operations to be disclosed in theembodiments of the present invention may be changed. Some components orcharacteristics of any embodiment may also be included in otherembodiments, or may be replaced with those of the other embodiments asnecessary.

In describing the present invention, if it is determined that thedetailed description of a related known function or construction rendersthe scope of the present invention unnecessarily ambiguous, the detaileddescription thereof will be omitted.

In the entire specification, when a certain portion “comprises orincludes” a certain component, this indicates that the other componentsare not excluded and may be further included unless specially describedotherwise. The terms “unit”, “-or/er” and “module” described in thespecification indicate a unit for processing at least one function oroperation, which may be implemented by hardware, software or acombination thereof. The words “a or an”, “one”, “the” and words relatedthereto may be used to include both a singular expression and a pluralexpression unless the context describing the present invention(particularly, the context of the following claims) clearly indicatesotherwise.

In this document, the embodiments of the present invention have beendescribed centering on a data transmission and reception relationshipbetween a mobile station and a base station. The base station may mean aterminal node of a network which directly performs communication with amobile station. In this document, a specific operation described asperformed by the base station may be performed by an upper node of thebase station.

Namely, it is apparent that, in a network comprised of a plurality ofnetwork nodes including a base station, various operations performed forcommunication with a mobile station may be performed by the basestation, or network nodes other than the base station. The term basestation may be replaced with the terms fixed station, Node B, eNode B(eNB), advanced base station (ABS), access point, etc.

The term mobile station (MS) may be replaced with user equipment (UE),subscriber station (SS), mobile subscriber station (MSS), mobileterminal, advanced mobile station (AMS), terminal, etc.

A transmitter refers to a fixed and/or mobile node for transmitting adata or voice service and a receiver refers to a fixed and/or mobilenode for receiving a data or voice service. Accordingly, in uplink, amobile station becomes a transmitter and a base station becomes areceiver. Similarly, in downlink transmission, a mobile station becomesa receiver and a base station becomes a transmitter.

Communication of a device with a “cell” may mean that the devicetransmit and receive a signal to and from a base station of the cell.That is, although a device substantially transmits and receives a signalto a specific base station, for convenience of description, anexpression “transmission and reception of a signal to and from a cellformed by the specific base station” may be used. Similarly, the term“macro cell” and/or “small cell” may mean not only specific coverage butalso a “macro base station supporting the macro cell” and/or a “smallcell base station supporting the small cell”.

The embodiments of the present invention can be supported by thestandard documents disclosed in any one of wireless access systems, suchas an IEEE 802.xx system, a 3rd Generation Partnership Project (3GPP)system, a 3GPP Long Term Evolution (LTE) system, and a 3GPP2 system.That is, the steps or portions, which are not described in order to makethe technical spirit of the present invention clear, may be supported bythe above documents.

In addition, all the terms disclosed in the present document may bedescribed by the above standard documents. In particular, theembodiments of the present invention may be supported by at least one ofP802.16-2004, P802.16e-2005, P802.16.1, P802.16p and P802.16.1bdocuments, which are the standard documents of the IEEE 802.16 system.

Hereinafter, the preferred embodiments of the present invention will bedescribed with reference to the accompanying drawings. It is to beunderstood that the detailed description which will be disclosed alongwith the accompanying drawings is intended to describe the exemplaryembodiments of the present invention, and is not intended to describe aunique embodiment which the present invention can be carried out.

It should be noted that specific terms disclosed in the presentinvention are proposed for convenience of description and betterunderstanding of the present invention, and the use of these specificterms may be changed to another format within the technical scope orspirit of the present invention.

1. IEEE 802.11 System Overview

1.1 Structure of WLAN System

FIG. 1 is a diagram showing an exemplary structure of an IEEE 802.11system to which the present invention is applicable.

An IEEE 802.11 structure may be composed of a plurality of componentsand a wireless local area network (WLAN) supporting station (STA)mobility transparent to a higher layer may be provided by interactionamong the components. A basic service set (BSS) may correspond to abasic component block in an IEEE 802.11 LAN. In FIG. 1, two BSSs (BSS1and BSS2) are present and each BSS includes two STAs (STA1 and STA2 areincluded in BSS1 and STA3 and STA4 are included in BSS2) as members. InFIG. 1, an ellipse indicating the BSS indicates a coverage area in whichSTAs included in the BSS maintains communication. This area may bereferred to as a basic service area (BSA). If an STA moves out of a BSA,the STA cannot directly communicate with other STAs in the BSA.

In an IEEE 802.11 LAN, a BSS is basically an independent BSS (IBSS). Forexample, the IBSS may have only two STAs. In addition, the simplest BSS(BSS1 or BSS2) of FIG. 1, in which other components are omitted, maycorrespond to a representative example of the IBSS. Such a configurationis possible when STAs can directly perform communication. In addition,such a LAN is not configured in advance but may be configured if a LANis necessary. This LAN may also be referred to as an ad-hoc network.

If an STA is turned on or off or if an STA enters or moves out of a BSS,the membership of the STA in the BSS may be dynamically changed. An STAmay join a BSS using a synchronization process in order to become amember of the BSS. In order to access all services of a BSS basedstructure, an STA should be associated with the BSS. Such associationmay be dynamically set and may include use of a distribution systemservice (DSS).

FIG. 2 is a diagram showing another exemplary structure of an IEEE802.11 system to which the present invention is applicable. In FIG. 2, adistribution system (DS), a distribution system medium (DSM) and anaccess point (AP) are added to the structure of FIG. 1.

In a LAN, a direct station-to-station distance may be restricted by PHYperformance Although such distance restriction may be possible,communication between stations located at a longer distance may benecessary. In order to support extended coverage, a DS may beconfigured.

The DS means a structure in which BSSs are mutually connected. Morespecifically, the BSSs are not independently present as shown in FIG. 1but the BSS may be present as an extended component of a networkincluding a plurality of BSSs.

The DS is a logical concept and may be specified by characteristics ofthe DSM. In IEEE 802.11 standards, a wireless medium (WM) and a DSM arelogically distinguished. Logical media are used for different purposesand are used by different components. In IEEE 802.11 standards, suchmedia are not restricted to the same or different media. Since pluralmedia are logically different, an IEEE 802.11 LAN structure (a DSstructure or another network structure) may be flexible. That is, theIEEE 802.11 LAN structure may be variously implemented and a LANstructure may be independently specified by physical properties of eachimplementation.

The DS provides seamless integration of a plurality of BSSs and provideslogical services necessary to treat an address to a destination so as tosupport a mobile apparatus.

The AP means an entity which enables associated STAs to access the DSvia the WM and has STA functionality. Data transfer between the BSS andthe DS may be performed via the AP. For example, STA2 and STA3 shown inFIG. 2 have STA functionality and provide a function enabling associatedSTAs (STA1 and STA4) to access the DS. In addition, since all APscorrespond to STAs, all APs may be addressable entities. An address usedby the AP for communication on the WM and an address used by the AP forcommunication on the DSM may not be equal.

Data transmitted from one of STAs associated with the AP to the STAaddress of the AP may always be received by an uncontrolled port andprocessed by an IEEE 802.1X port access entity. In addition, if acontrolled port is authenticated, transmission data (or frames) may betransmitted to the DS.

FIG. 3 is a diagram showing another exemplary structure of an IEEE802.11 system to which the present invention is applicable. In FIG. 3,an extended service set (ESS) for providing wide coverage is added tothe structure of FIG. 2.

A wireless network having an arbitrary size and complexity may becomposed of a DS and BSSs. In an IEEE 802.11 system, such a network isreferred to as an ESS network. The ESS may correspond to a set of BSSsconnected to one DS. However, the ESS does not include the DS. The ESSnetwork appears as an IBSS network at a logical link control (LLC)layer. STAs included in the ESS may communicate with each other andmobile STAs may move from one BSS to another BSS (within the same ESS)transparently to the LLC layer.

In IEEE 802.11, relative physical locations of the BSSs in FIG. 3 arenot assumed and may be defined as follows. The BSSs may partiallyoverlap in order to provide consecutive coverage. In addition, the BSSsmay not be physically connected and a distance between BSSs is notlogically restricted. In addition, the BSSs may be physically located atthe same location in order to provide redundancy. In addition, one (ormore) IBSS or ESS network may be physically present in the same space asone (or more) ESS network. This corresponds to an ESS network type suchas a case in which an ad-hoc network operates at a location where theESS network is present, a case in which IEEE 802.11 networks physicallyoverlapped by different organizations are configured or a case in whichtwo or more different access and security policies are necessary at thesame location.

FIG. 4 is a diagram showing an exemplary structure of a WLAN system.FIG. 4 shows an example of an infrastructure BSS including a DS.

In the example of FIG. 4, BSS1 and BSS2 configure an ESS. In the WLANsystem, an STA operates according to a MAC/PHY rule of IEEE 802.11. TheSTA includes an AP STA and a non-AP STA. The non-AP STA corresponds toan apparatus directly handled by a user, such as a laptop or a mobilephone. In the example of FIG. 4, STA1, STA3 and STA4 correspond to thenon-AP STA and STA2 and STA5 correspond to the AP STA.

In the following description, the non-AP STA may be referred to as aterminal, a wireless transmit/receive unit (WTRU), a user equipment(UE), a mobile station (MS), a mobile terminal or a mobile subscriberstation (MSS). In addition, the AP may correspond to a base station(BS), a Node-B, an evolved Node-B (eNB), a base transceiver system (BTS)or a femto BS.

1.2 Link Setup Process

FIG. 5 is a diagram illustrating a general link setup process.

In order to establish a link with respect to a network and perform datatransmission and reception, an STA discovers the network, performsauthentication, establishes association and performs an authenticationprocess for security. The link setup process may be referred to as asession initiation process or a session setup process. In addition,discovery, authentication, association and security setup of the linksetup process may be collectively referred to as an association process.

An exemplary link setup process will be described with reference to FIG.5.

In step S510, the STA may perform a network discovery operation. Thenetwork discovery operation may include a scanning operation of the STA.That is, the STA discovers the network in order to access the network.The STA should identify a compatible network before participating in awireless network and a process of identifying a network present in aspecific area is referred to as scanning. The scanning method includesan active scanning method and a passive scanning method.

In FIG. 5, a network discovery operation including an active scanningprocess is shown. In active scanning, the STA which performs scanningtransmits a probe request frame while moving between channels and waitsfor a response thereto, in order to detect which AP is present. Aresponder transmits a probe response frame to the STA, which transmittedthe probe request frame, as a response to the probe request frame. Theresponder may be an STA which lastly transmitted a beacon frame in a BSSof a scanned channel. In the BSS, since the AP transmits the beaconframe, the AP is the responder. In the IBSS, since the STAs in the IBSSalternately transmit the beacon frame, the responder is not fixed. Forexample, the STA which transmits the probe request frame on a firstchannel and receives the probe response frame on the first channelstores BSS related information included in the received probe responseframe, moves to a next channel (e.g., a second channel) and performsscanning (probe request/response transmission/reception on the secondchannel) using the same method.

Although not shown in FIG. 5, a scanning operation may be performedusing a passive scanning method. In passive scanning, the STA whichperforms scanning waits for a beacon frame while moving betweenchannels. The beacon frame is a management frame in IEEE 802.11 and isperiodically transmitted in order to indicate presence of a wirelessnetwork and to enable the STA, which performs scanning, to discover andparticipate in the wireless network. In the BSS, the AP is responsiblefor periodically transmitting the beacon frame. In the IBSS, the STAsalternately transmit the beacon frame. The STA which performs scanningreceives the beacon frame, stores information about the BSS included inthe beacon frame, and records beacon frame information of each channelwhile moving to another channel. The STA which receives the beacon framemay store BSS related information included in the received beacon frame,move to a next channel and perform scanning on the next channel usingthe same method.

Active scanning has delay and power consumption less than those ofpassive scanning.

After the STA has discovered the network, an authentication process maybe performed in step S520. Such an authentication process may bereferred to as a first authentication process to be distinguished from asecurity setup operation of step S540.

The authentication process includes a process of, at the STA,transmitting an authentication request frame to the AP and, at the AP,transmitting an authentication response frame to the STA in responsethereto. The authentication frame used for authenticationrequest/response corresponds to a management frame.

The authentication frame may include information about an authenticationalgorithm number, an authentication transaction sequence number, astatus code, a challenge text, a robust security network (RSN), a finitecyclic group, etc. The information may be examples of informationincluded in the authentication request/response frame and may bereplaced with other information. The information may further includeadditional information.

The STA may transmit the authentication request frame to the AP. The APmay determine whether authentication of the STA is allowed, based on theinformation included in the received authentication request frame. TheAP may provide the STA with the authentication result via theauthentication response frame.

After the STA is successfully authenticated, an association process maybe performed in step S530. The association process includes a processof, at the STA, transmitting an association request frame to the AP and,at the AP, transmitting an association response frame to the STA inresponse thereto.

For example, the association request frame may include information aboutvarious capabilities, beacon listen interval, service set identifier(SSID), supported rates, RSN, mobility domain, supported operatingclasses, traffic indication map (TIM) broadcast request, interworkingservice capability, etc.

For example, the association response frame may include informationabout various capabilities, status code, association ID (AID), supportedrates, enhanced distributed channel access (EDCA) parameter set,received channel power indicator (RCPI), received signal to noiseindicator (RSNI), mobility domain, timeout interval (associationcomeback time), overlapping BSS scan parameter, TIM broadcast response,QoS map, etc.

This information is purely exemplary information included in theassociation request/response frame and may be replaced with otherinformation. This information may further include additionalinformation.

After the STA is successfully authenticated, a security setup processmay be performed in step S540. The security setup process of step S540may be referred to as an authentication process through a robustsecurity network association (RSNA) request/response. The authenticationprocess of step S520 may be referred to as the first authenticationprocess and the security setup process of step S540 may be simplyreferred to as an authentication process.

The security setup process of step S540 may include a private key setupprocess through 4-way handshaking of an extensible authenticationprotocol over LAN (EAPOL) frame. In addition, the security setup processmay be performed according to a security method which is not defined inthe IEEE 802.11 standard.

2.1 Evolution of WLAN

As a technical standard recently established in order to overcomelimitations in communication speed in a WLAN, IEEE 802.11n has beendevised. IEEE 802.11n aims at increasing network speed and reliabilityand extending wireless network distance. More specifically, IEEE 802.11nis based on multiple input and multiple output (MIMO) technology usingmultiple antennas in a transmitter and a receiver in order to supporthigh throughput (HT) with a maximum data rate of 540 Mbps or more, tominimize transmission errors, and to optimize data rate.

As WLANs have come into widespread use and applications using the samehave been diversified, recently, there is a need for a new WLAN systemsupporting throughput higher than a data rate supported by IEEE 802.11n.A next-generation WLAN system supporting very high throughput (VHT) is anext version (e.g., IEEE 802.11ac) of the IEEE 802.11n WLAN system andis an IEEE 802.11 WLAN system newly proposed in order to support a datarate of 1 Gbps or more at a MAC service access point (SAP).

The next-generation WLAN system supports a multi-user MIMO (MU-MIMO)transmission scheme by which a plurality of STAs simultaneously accessesa channel in order to efficiently use a radio channel. According to theMU-MIMO transmission scheme, the AP may simultaneously transmit packetsto one or more MIMO-paired STAs.

In addition, support of a WLAN system operation in a whitespace is beingdiscussed. For example, introduction of a WLAN system in a TV whitespace(WS) such as a frequency band (e.g., 54 to 698 MHz) in an idle state dueto digitalization of analog TVs is being discussed as the IEEE 802.11afstandard. However, this is only exemplary and the whitespace may beincumbently used by a licensed user. The licensed user means a user whois allowed to use a licensed band and may be referred to as a licenseddevice, a primary user or an incumbent user.

For example, the AP and/or the STA which operate in the WS shouldprovide a protection function to the licensed user. For example, if alicensed user such as a microphone already uses a specific WS channelwhich is a frequency band divided on regulation such that a WS band hasa specific bandwidth, the AP and/or the STA cannot use the frequencyband corresponding to the WS channel in order to protect the licenseduser. In addition, the AP and/or the STA must stop use of the frequencyband if the licensed user uses the frequency band used for transmissionand/or reception of a current frame.

Accordingly, the AP and/or the STA should perform a procedure ofdetermining whether a specific frequency band in a WS band is available,that is, whether a licensed user uses the frequency band. Determiningwhether a licensed user uses a specific frequency band is referred to asspectrum sensing. As a spectrum sensing mechanism, an energy detectionmethod, a signature detection method, etc. may be used. It may bedetermined that the licensed user uses the frequency band if receivedsignal strength is equal to or greater than a predetermined value or ifa DTV preamble is detected.

In addition, as next-generation communication technology,machine-to-machine (M2M) communication technology is being discussed.Even in an IEEE 802.11 WLAN system, a technical standard supporting M2Mcommunication has been developed as IEEE 802.11ah. M2M communicationmeans a communication scheme including one or more machines and may bereferred to as machine type communication (MTC). Here, a machine meansan entity which does not require direct operation or intervention of aperson. For example, a device including a mobile communication module,such as a meter or a vending machine, may include a user equipment suchas a smart phone which is capable of automatically accessing a networkwithout operation/intervention of a user to perform communication. M2Mcommunication includes communication between devices (e.g.,device-to-device (D2D) communication) and communication between a deviceand an application server. Examples of communication between a deviceand a server include communication between a vending machine and aserver, communication between a point of sale (POS) device and a serverand communication between an electric meter, a gas meter or a watermeter and a server. An M2M communication based application may includesecurity, transportation, health care, etc. If the characteristics ofsuch examples are considered, in general, M2M communication shouldsupport transmission and reception of a small amount of data at a lowrate in an environment in which very many apparatuses are present.

More specifically, M2M communication should support a larger number ofSTAs. In a currently defined WLAN system, it is assumed that a maximumof 2007 STAs is associated with one AP. However, in M2M communication,methods supporting the case in which a larger number of STAs (about6000) are associated with one AP are being discussed. In addition, inM2M communication, it is estimated that there are many applicationssupporting/requiring a low transfer rate. In order to appropriatelysupport the low transfer rate, for example, in a WLAN system, the STAmay recognize presence of data to be transmitted thereto based on atraffic indication map (TIM) element and methods of reducing a bitmapsize of the TIM are being discussed. In addition, in M2M communication,it is estimated that there is traffic having a very longtransmission/reception interval. For example, in electricity/gas/waterconsumption, a very small amount of data is required to be exchanged ata long period (e.g., one month). In a WLAN system, although the numberof STAs associated with one AP is increased, methods of efficientlysupporting the case in which the number of STAs, in which a data frameto be received from the AP is present during one beacon period, is verysmall are being discussed.

WLAN technology has rapidly evolved. In addition to the above-describedexamples, technology for direct link setup, improvement of mediastreaming performance, support of fast and/or large-scale initialsession setup, support of extended bandwidth and operating frequency,etc. is being developed.

2.2 Medium Access Mechanism

In a WLAN system according to IEEE 802.11, the basic access mechanism ofmedium access control (MAC) is a carrier sense multiple access withcollision avoidance (CSMA/CA) mechanism. The CSMA/CA mechanism is alsoreferred to as a distributed coordination function (DCF) of IEEE 802.11MAC and employs a “listen before talk” access mechanism. According tosuch an access mechanism, the AP and/or the STA may perform clearchannel assessment (CCA) for sensing a radio channel or medium during apredetermined time interval (for example, a DCF inter-frame space(DIFS)) before starting transmission. If it is determined that themedium is in an idle state as the sensed result, frame transmissionstarts via the medium. If it is determined that the medium is in anoccupied state, the AP and/or the STA may set and wait for a delayperiod (e.g., a random backoff period) for medium access withoutstarting transmission and then attempt to perform frame transmission.Since several STAs attempt to perform frame transmission after waitingfor different times by applying the random backoff period, it ispossible to minimize collision.

In addition, the IEEE 802.11 MAC protocol provides a hybrid coordinationfunction (HCF). The HCF is based on the DCF and a point coordinationfunction (PCF). The PCF refers to a periodic polling method for enablingall reception AP and/or STAs to receive data frames using a pollingbased synchronous access method. In addition, the HCF has enhanceddistributed channel access (EDCA) and HCF controlled channel access(HCCA). The EDCA uses a contention access method for providing dataframes to a plurality of users by a provider and the HCCA uses acontention-free channel access method using a polling mechanism. Inaddition, the HCF includes a medium access mechanism for improvingquality of service (QoS) of a WLAN and may transmit QoS data both in acontention period (CP) and a contention free period (CFP).

FIG. 6 is a diagram illustrating a backoff process.

Operation based on a random backoff period will be described withreference to FIG. 6. If a medium is changed from an occupied or busystate to an idle state, several STAs may attempt data (or frame)transmission. At this time, a method of minimizing collision, the STAsmay select respective random backoff counts, wait for slot timescorresponding to the random backoff counts and attempt transmission. Therandom backoff count has a pseudo-random integer and may be set to oneof values of 0 to CW. Here, the CW is a contention window parametervalue. The CW parameter is set to CWmin as an initial value but may beset to twice CWmin if transmission fails (e.g., ACK for the transmissionframe is not received). If the CW parameter value becomes CWmax, datatransmission may be attempted while maintaining the CWmax value untildata transmission is successful. If data transmission is successful, theCW parameter value is reset to CWmin. CW, CWmin and CWmax values arepreferably set to 2n−1 (n=0, 1, 2, . . . ).

If the random backoff process starts, the STA continuously monitors themedium while the backoff slots are counted down according to the setbackoff count value. If the medium is in the occupied state, countdownis stopped and, if the medium is in the idle state, countdown isresumed.

In the example of FIG. 6, if packets to be transmitted to the MAC ofSTA3 arrive, STA3 may confirm that the medium is in the idle stateduring the DIFS and immediately transmit a frame. Meanwhile, theremaining STAs monitor that the medium is in the busy state and wait.During a wait time, data to be transmitted may be generated in STA1,STA2 and STA5. The STAs may wait for the DIFS if the medium is in theidle state and then count down the backoff slots according to therespectively selected random backoff count values.

In the example of FIG. 6, STA2 selects a smallest backoff count valueand STA1 selects a largest backoff count value. That is, the residualbackoff time of STA5 is less than the residual backoff time of STA1 whenSTA2 completes backoff count and starts frame transmission. STA1 andSTA5 stop countdown and wait while STA2 occupies the medium. Ifoccupancy of the medium by STA2 ends and the medium enters the idlestate again, STA1 and STA5 wait for the DIFS and then resume countdown.That is, after the residual backoff slots corresponding to the residualbackoff time are counted down, frame transmission may start. Since theresidual backoff time of STA5 is less than of STA1, STA5 starts frametransmission.

If STA2 occupies the medium, data to be transmitted may be generated inthe STA4. At this time, STA4 may wait for the DIFS if the medium entersthe idle state, perform countdown according to a random backoff countvalue selected thereby, and start frame transmission. In the example ofFIG. 6, the residual backoff time of STA5 accidentally matches therandom backoff time of STA4. In this case, collision may occur betweenSTA4 and STA5. If collision occurs, both STA4 and STA5 do not receiveACK and data transmission fails. In this case, STA4 and STA5 may doublethe CW value, select the respective random backoff count values and thenperform countdown. STA1 may wait while the medium is busy due totransmission of STA4 and STA5, wait for the DIFS if the medium entersthe idle state, and start frame transmission if the residual backofftime has elapsed.

2.3 Sensing Operation of STA

As described above, the CSMA/CA mechanism includes not only physicalcarrier sensing for directly sensing a medium by an AP and/or an STA butalso virtual carrier sensing. Virtual carrier sensing solves a problemwhich may occur in medium access, such as a hidden node problem. Forvirtual carrier sensing, MAC of a WLAN may use a network allocationvector (NAV). The NAV refers to a value of a time until a medium becomesavailable, which is indicated to another AP and/or STA by an AP and/oran STA, which is currently utilizing the medium or has rights to utilizethe medium. Accordingly, the NAV value corresponds to a period of timewhen the medium will be used by the AP and/or the STA for transmittingthe frame, and medium access of the STA which receives the NAV value isprohibited during that period of time. The NAV may be set according tothe value of the “duration” field of a MAC header of a frame.

A robust collision detection mechanism for reducing collision has beenintroduced, which will be described with reference to FIGS. 7 and 8.Although a transmission range may not be equal to an actual carriersensing range, for convenience, assume that the transmission range maybe equal to the actual carrier sensing range.

FIG. 7 is a diagram illustrating a hidden node and an exposed node.

FIG. 7(a) shows a hidden node, and, in this case, an STA A and an STA Bare performing communication and an STA C has information to betransmitted. More specifically, although the STA A transmits informationto the STA B, the STA C may determine that the medium is in the idlestate, when carrier sensing is performed before transmitting data to theSTA B. This is because the STA C may not sense transmission of the STA A(that is, the medium is busy). In this case, since the STA Bsimultaneously receives information of the STA A and the STA C,collision occurs. At this time, the STA A may be the hidden node of theSTA C.

FIG. 7(b) shows an exposed node and, in this case, the STA B transmitsdata to the STA A and the STA C has information to be transmitted to theSTA D. In this case, if the STA C performs carrier sensing, it may bedetermined that the medium is busy due to transmission of the STA B. Ifthe STA C has information to be transmitted to the STA D, since it issensed that the medium is busy, the STA C waits until the medium entersthe idle state. However, since the STA A is actually outside thetransmission range of the STA C, transmission from the STA C andtransmission from the STA B may not collide from the viewpoint of theSTA A. Therefore, the STA C unnecessarily waits until transmission ofthe STA B is stopped. At this time, the STA C may be the exposed node ofthe STA B.

FIG. 8 is a diagram illustrating request to send (RTS) and clear to send(CTS).

In the example of FIG. 7, in order to efficiently use a collisionavoidance mechanism, short signaling packet such as RTS and CTS may beused. RST/CTS between two STAs may be enabled to be overheard byperipheral STAs such that the peripheral STAs confirm informationtransmission between the two STAs. For example, if a transmission STAtransmits an RTS frame to a reception STA, the reception STA transmits aCTS frame to peripheral UEs to inform the peripheral UEs that thereception STA receives data.

FIG. 8(a) shows a method of solving a hidden node problem. Assume thatboth the STA A and the STA C attempt to transmit data to the STA B. Ifthe STA A transmits the RTS to the STA B, the STA B transmits the CTS tothe peripheral STA A and C. As a result, the STA C waits until datatransmission of the STA A and the STA B is finished, thereby avoidingcollision.

FIG. 8(b) shows a method of solving an exposed node problem. The STA Cmay overhear RTS/CTS transmission between the STA A and the STA B anddetermine that collision does not occur even when the STA C transmitsdata to another STA (e.g., the STA D). That is, the STA B transmits theRTS to all peripheral UEs and transmits the CTS only to the STA A havingdata to be actually transmitted. Since the STA C receives the RTS butdoes not receive the CTS of the STA A, it can be confirmed that the STAA is outside carrier sensing of the STA C.

2.4 Power Management

As described above, in a WLAN system, channel sensing should beperformed before an STA performs transmission and reception. When thechannel is always sensed, continuous power consumption of the STA iscaused. Power consumption in a reception state is not substantiallydifferent from power consumption in a transmission state andcontinuously maintaining the reception state imposes a burden on an STAwith limited power (that is, operated by a battery). Accordingly, if areception standby state is maintained such that the STA continuouslysenses the channel, power is inefficiently consumed without any specialadvantage in terms of WLAN throughput. In order to solve such a problem,in a WLAN system, a power management (PM) mode of the STA is supported.

The PM mode of the STA is divided into an active mode and a power save(PS) mode. The STA fundamentally operates in an active mode. The STAwhich operates in the active mode is maintained in an awake state. Theawake state refers to a state in which normal operation such as frametransmission and reception or channel scanning is possible. The STAwhich operates in the PS mode operates while switching between a sleepstate or an awake state. The STA which operates in the sleep stateoperates with minimum power and does not perform frame transmission andreception or channel scanning.

Since power consumption is reduced as the sleep state of the STA isincreased, the operation period of the STA is increased. However, sinceframe transmission and reception is impossible in the sleep state, theSTA may not unconditionally operate in the sleep state. If a frame to betransmitted from the STA, which operates in the sleep state, to the APis present, the STA may be switched to the awake state to transmit theframe. If a frame to be transmitted from the AP to the STA is present,the STA in the sleep state may not receive the frame and may not confirmthat the frame to be received is present. Accordingly, the STA needs toperform an operation for switching to the awake state according to aspecific period in order to confirm presence of the frame to betransmitted thereto (to receive the frame if the frame to be transmittedis present).

FIG. 9 is a diagram illustrating power management operation.

Referring to FIG. 9, an AP 210 transmits beacon frames to STAs within aBSS at a predetermined period (S211, S212, S213, S214, S215 and S216).The beacon frame includes a traffic indication map (TIM) informationelement. The TIM information element includes information indicatingthat buffered traffic for STAs associated with the AP 210 is present andthe AP 210 will transmit a frame. The TIM element includes a TIM used toindicate a unicast frame or a delivery traffic indication map (DTIM)used to indicate a multicast or broadcast frame.

The AP 210 may transmit the DTIM once whenever the beacon frame istransmitted three times. An STA1 220 and an STA2 222 operate in the PSmode. The STA1 220 and the STA2 222 may be switched from the sleep stateto the awake state at a predetermined wakeup interval to receive a TIMelement transmitted by the AP 210. Each STA may compute a time to switchto the awake state based on a local clock thereof. In the example ofFIG. 9, assume that the clock of the STA matches the clock of the AP.

For example, the predetermined awake interval may be set such that theSTA1 220 is switched to the awake state every beacon interval to receivea TIM element. Accordingly, the STA1 220 may be switched to the awakestate (S211) when the AP 210 first transmits the beacon frame (S211).The STA1 220 may receive the beacon frame and acquire the TIM element.If the acquired TIM element indicates that a frame to be transmitted tothe STA1 220 is present, the STA1 220 may transmit, to the AP 210, apower save-Poll (PS-Poll) frame for requesting frame transmission fromthe AP 210 (S221 a). The AP 210 may transmit the frame to the STA1 220in correspondence with the PS-Poll frame (S231). The STA1 220 whichcompletes frame reception is switched to the sleep state.

When the AP 210 secondly transmits the beacon frame, since anotherdevice access the medium and thus the medium is busy, the AP 210 may nottransmit the beacon frame at an accurate beacon interval and maytransmit the beacon frame at a delayed time (S212). In this case, theoperation mode of the STA1 220 is switched to the awake state accordingto the beacon interval but the delayed beacon frame is not received.Therefore, the operation mode of the STA1 220 is switched to the sleepstate again (S222).

When the AP 210 thirdly transmits the beacon frame, the beacon frame mayinclude a TIM element set to a DTIM. Since the medium is busy, the AP210 transmits the beacon frame at a delayed time (S213). The STA1 220 isswitched to the awake state according to the beacon interval and mayacquire the DTIM via the beacon frame transmitted by the AP 210. Assumethat the DTIM acquired by the STA1 220 indicates that a frame to betransmitted to the STA1 220 is not present and a frame for another STAis present. In this case, the STA1 220 may confirm that a frametransmitted thereby is not present and may be switched to the sleepstate again. The AP 210 transmits the beacon frame and then transmitsthe frame to the STA (S232).

The AP 210 fourthly transmits the beacon frame (S214). Since the STA1220 cannot acquire information indicating that buffered traffic thereforis present via reception of the TIM element twice, the wakeup intervalfor receiving the TIM element may be controlled. Alternatively, ifsignaling information for controlling the wakeup interval of the STA1220 is included in the beacon frame transmitted by the AP 210, thewakeup interval value of the STA1 220 may be controlled. In the presentexample, the STA1 220 may change switching of the operation state forreceiving the TIM element every beacon interval to switching of theoperation state every three beacon intervals. Accordingly, since theSTA1 220 is maintained in the sleep state when the AP 210 transmits thefourth beacon frame (S214) and transmits the fifth beacon frame (S215),the TIM element cannot be acquired.

When the AP 210 sixthly transmits the beacon frame (S216), the STA1 220may be switched to the awake state to acquire the TIM element includedin the beacon frame (S224). Since the TIM element is a DTIM indicatingthat a broadcast frame is present, the STA1 220 may not transmit thePS-Poll frame to the AP 210 but may receive a broadcast frametransmitted by the AP 210 (S234). The wakeup interval set in the STA2230 may be set to be greater than that of the STA1 220. Accordingly, theSTA2 230 may be switched to the awake state to receive the TIM element(S241), when the AP 210 fifthly transmits the beacon frame (S215). TheSTA2 230 may confirm that a frame to be transmitted thereto is presentvia the TIM element and transmits the PS-Poll frame to the AP 210 (S241a) in order to request frame transmission. The AP 210 may transmit theframe to the STA2 230 in correspondence with the PS-Poll frame (S233).

For PM management shown in FIG. 9, a TIM element includes a TIMindicating whether a frame to be transmitted to an STA is present and aDTIM indicating whether a broadcast/multicast frame is present. The DTIMmay be implemented by setting a field of the TIM element.

FIGS. 10 to 12 are diagrams illustrating operation of a station (STA)which receives a traffic indication map (TIM).

Referring to FIG. 10, an STA may be switched from a sleep state to anawake state in order to receive a beacon frame including a TIM from anAP and interpret the received TIM element to confirm that bufferedtraffic to be transmitted thereto is present. The STA may contend withother STAs for medium access for transmitting a PS-Poll frame and thentransmit the PS-Poll frame in order to request data frame transmissionfrom the AP. The AP which receives the PS-Poll frame transmitted by theSTA may transmit the frame to the STA. The STA may receive the dataframe and transmit an ACK frame to the AP. Thereafter, the STA may beswitched to the sleep state again.

As shown in FIG. 10, the AP may receive the PS-Poll frame from the STAand then operate according to an immediate response method fortransmitting a data frame after a predetermined time (e.g., a shortinter-frame space (SIFS)). If the AP does not prepare a data frame to betransmitted to the STA during the SIFS after receiving the PS-Pollframe, the AP may operate according to a deferred response method, whichwill be described with reference to FIG. 11.

In the example of FIG. 11, operation for switching the STA from thesleep state to the awake state, receiving a TIM from the AP, contendingand transmitting a PS-Poll frame to the AP is equal to that of FIG. 10.If the data frame is not prepared during the SIFS even when the APreceives the PS-Poll frame, the data frame is not transmitted but an ACKframe may be transmitted to the STA. If the data frame is prepared aftertransmitting the ACK frame, the AP may contend and transmit the dataframe to the STA. The STA may transmit the ACK frame indicating that thedata frame has been successfully received to the AP and may be switchedto the sleep state.

FIG. 12 shows an example in which the AP transmits the DTIM. The STAsmay be switched from the sleep state to the awake state in order toreceive the beacon frame including the DTIM element from the AP. The STAmay confirm that a multicast/broadcast frame will be transmitted via thereceived DTIM. The AP may immediately transmit data (that is, amulticast/broadcast frame) without PS-Poll frame transmission andreception after transmitting the beacon frame including the DTIM. TheSTAs may receive data in the awake state after receiving the beaconframe including the DTIM and may be switched to the sleep state againafter completing data reception.

2.5 TIM Structure

In the PM mode management method based on the TIM (or DTIM) protocoldescribed with reference to FIGS. 9 to 12, the STAs may confirm whethera data frame to be transmitted thereto is present via STA identificationincluded in the TIM element. The STA identification may be related to anassociation identifier (AID) assigned to the STA upon association withthe AP.

The AID is used as a unique identifier for each STA within one BSS. Forexample, in a current WLAN system, the AID may be one of values of 1 to2007. In a currently defined WLAN system, 14 bits are assigned to theAID in a frame transmitted by the AP and/or the STA. Although up to16383 may be assigned as the AID value, 2008 to 16383 may be reserved.

The TIM element according to an existing definition is not appropriatelyapplied to an M2M application in which a large number (e.g., more than2007) of STAs is associated with one AP. If the existing TIM structureextends without change, the size of the TIM bitmap is too large to besupported in an existing frame format and to be suitable for M2Mcommunication considering an application with a low transfer rate. Inaddition, in M2M communication, it is predicted that the number of STAs,in which a reception data frame is present during one beacon period, isvery small. Accordingly, in M2M communication, since the size of the TIMbitmap is increased but most bits have a value of 0, there is a need fortechnology for efficiently compressing the bitmap.

As an existing bitmap compression technology, a method of omitting 0which continuously appears at a front part of a bitmap and defining anoffset (or a start point) is provided. However, if the number of STAs inwhich a buffered frame is present is small but a difference between theAID values of the STAs is large, compression efficiency is bad. Forexample, if only frames to be transmitted to only two STAs respectivelyhaving AID values of 10 and 2000 are buffered, the length of thecompressed bitmap is 1990 but all bits other than both ends have a valueof 0. If the number of STAs which may be associated with one AP issmall, bitmap compression inefficiency is not problematic but, if thenumber of STAs is increased, bitmap compression inefficiencydeteriorates overall system performance.

As a method of solving this problem, AIDs may be divided into severalgroups to more efficiently perform data transmission. A specific groupID (GID) is assigned to each group. AIDs assigned based on the groupwill be described with reference to FIG. 13.

FIG. 13(a) shows an example of AIDs assigned based on a group. In theexample of FIG. 13(a), several bits of a front part of the AID bitmapmay be used to indicate the GID. For example, four DIDs may be expressedby the first two bits of the AID of the AID bitmap. If the total lengthof the AID bitmap is N bits, the first two bits (B1 and B2) indicate theGID of the AID.

FIG. 13(a) shows another example of AIDs assigned based on a group. Inthe example of FIG. 13(b), the GID may be assigned according to thelocation of the AID. At this time, the AIDs using the same GID may beexpressed by an offset and a length value. For example, if GID 1 isexpressed by an offset A and a length B, this means that AIDs of A toA+B−1 on the bitmap have GID 1. For example, in the example of FIG.13(b), assume that all AIDs of 1 to N4 are divided into four groups. Inthis case, AIDs belonging to GID 1 are 1 to N1 and may be expressed byan offset 1 and a length N1. AIDs belonging to GID2 may be expressed byan offset N1+1 and a length N2−N1+1, AIDs belonging to GID 3 may beexpressed by an offset N2+1 and a length N3−N2+1, and AIDs belonging toGID 4 may be expressed by an offset N3+1 and a length N4−N3+1.

If the AIDs assigned based on the group are introduced, channel accessis allowed at a time interval which is changed according to the GID tosolve lack of TIM elements for a large number of STAs and to efficientlyperform data transmission and reception. For example, only channelaccess of STA(s) corresponding to a specific group may be granted duringa specific time interval and channel access of the remaining STA(s) maybe restricted. A predetermined time interval at which only access ofspecific STA(s) is granted may also be referred to as a restrictedaccess window (RAW).

Channel access according to GID will be described with reference to FIG.13(c). FIG. 13(c) shows a channel access mechanism according to a beaconinterval if the AIDs are divided into three groups. At a first beaconinterval (or a first RAW), channel access of STAs belonging to GID 1 isgranted but channel access of STAs belonging to other GIDs is notgranted. For such implementation, the first beacon includes a TIMelement for AIDs corresponding to GID 1. A second beacon frame includesa TIM element for AIDs corresponding to GID 2 and thus only channelaccess of the STAs corresponding to the AIDs belonging to GID 2 isgranted during the second beacon interval (or the second RAW). A thirdbeacon frame includes a TIM element for AIDs corresponding to GID 3 andthus only channel access of the STAs corresponding to the AIDs belongingto GID 3 is granted during the third beacon interval (or the third RAW).A fourth beacon frame includes a TIM element for AIDs corresponding toGID 1 and thus only channel access of the STAs corresponding to the AIDsbelonging to GID 1 is granted during the fourth beacon interval (or thefourth RAW). Only channel access of the STAs corresponding to a specificgroup indicated by the TIM included in the beacon frame may be grantedeven in fifth and subsequent beacon intervals (or fifth and subsequentRAWs).

Although the order of GIDs allowed according to the beacon interval iscyclic or periodic in FIG. 13(c), the present invention is not limitedthereto. That is, by including only AID(s) belonging to specific GID(s)in the TIM elements, only channel access of STA(s) corresponding to thespecific AID(s) may be granted during a specific time interval (e.g., aspecific RAW) and channel access of the remaining STA(s) may not begranted.

The above-described group based AID assignment method may also bereferred to as a hierarchical structure of a TIM. That is, an entire AIDspace may be divided into a plurality of blocks and only channel accessof STA(s) corresponding to a specific block having a non-zero value(that is, STAs of a specific group) may be granted. A TIM having a largesize is divided into small blocks/groups such that the STA easilymaintains TIM information and easily manages blocks/groups according toclass, QoS or usage of the STA. Although a 2-level layer is shown in theexample of FIG. 13, a TIM of a hierarchical structure having two or morelevels may be constructed. For example, the entire AID space may bedivided into a plurality of page groups, each page group may be dividedinto a plurality of blocks, and each block may be divided into aplurality of sub-blocks. In this case, as an extension of the example ofFIG. 13(a), the first N1 bits of the AID bitmap indicate a paging ID(that is, a PID), the next N2 bits indicate a block ID, the next N3 bitsindicate a sub-block ID, and the remaining bits indicate the STA bitlocation in the sub-block.

In the following examples of the present invention, various methods ofdividing and managing STAs (or AIDs assigned to the STAs) on apredetermined hierarchical group basis are applied and the group basedAID assignment method is not limited to the above examples.

2.6 Improved Channel Access Method

If AIDs are assigned/managed based on a group, STAs belonging to aspecific group may use a channel only at a “group channel accessinterval (or RAW)” assigned to the group. If an STA supports an M2Mapplication, traffic for the STA may have a property which may begenerated at a long period (e.g., several tens of minutes or severalhours). Since such an STA does not need to be in the awake statefrequently, the STA may be in the sleep mode for g a long period of timeand be occasionally switched to the awake state (that is, the awakeinterval of the STA may be set to be long). An STA having a long wakeupinterval may be referred to as an STA which operates in a “long-sleeper”or “long-sleep” mode. The case in which the wakeup interval is set to belong is not limited to M2M communication and the wakeup interval may beset to be long according to the state of the STA or surroundings of theSTA even in normal WLAN operation.

If the wakeup interval is set, the STA may determine whether a localclock thereof exceeds the wakeup interval. However, since the localclock of the STA generally uses a cheap oscillator, an error probabilityis high. In addition, if the STA operates in long-sleep mode, the errormay be increased with time. Accordingly, time synchronization of the STAwhich occasionally wakes up may not match time synchronization of theAP. For example, although the STA computes when the STA may receive thebeacon frame to be switched to the awake state, the STA may not actuallyreceive the beacon frame from the AP at that timing. That is, due toclock drift, the STA may miss the beacon frame and such a problem mayfrequently occur if the STA operates in the long sleep mode.

FIGS. 14 to 16 are diagrams showing examples of operation of an STA if agroup channel access interval is set.

In the example of FIG. 14, STA3 may belong to group 3 (that is, GID=3),wake up at a channel access interval assigned to group 1 and performPS-Poll for requesting frame transmission from the AP. The AP whichreceives PS-Poll from the STA transmits an ACK frame to STA3. Ifbuffered data to be transmitted to STA3 is present, the AP may provideinformation indicating that data to be transmitted is present via theACK frame. For example, the value of a “More Data” field (or an MDfield) having a size of 1 bit included in the ACK frame may be set to 1(that is, MD=1) to indicate the above information.

Since a time when STA3 transmits PS-Poll belongs to the channel accessinterval for group 1, even if data to be transmitted to STA3 is present,the AP does not immediately transmit data after transmitting the ACKframe but transmits data to STA3 at a channel access interval (GID 3channel access of FIG. 14) assigned to group 3 to which STA3 belongs.

Since STA3 receives the ACK frame set to MD=1 from the AP, STA3continuously waits for transmission of data from the AP. That is, in theexample of FIG. 14, since STA3 cannot receive the beacon frameimmediately after waking up, STA3 transmits PS-Poll to the AP on theassumption that a time when STA3 wakes up corresponds to the channelaccess interval assigned to the group, to which STA3 belongs, accordingto computation based on the local clock thereof and data to betransmitted thereto is present. Alternatively, since STA3 operates inthe long-sleep mode, on the assumption that time synchronization is notperformed, if the data to be transmitted thereto is present, STA3 maytransmit PS-Poll to the AP in order to receive the data. Since the ACKframe received by STA3 from the AP indicates that data to be transmittedto STA3 is present, STA3 continuously waits for data reception under theassumption of the interval in which channel access thereof is granted.STA3 unnecessarily consumes power even when data reception is notallowed, until time synchronization is appropriately performed frominformation included in a next beacon frame.

In particular, if STA3 operates in the long-sleep mode, the beacon framemay frequently not be received, CCA may be performed even at the channelaccess interval, to which STA2 does not belong, thereby causingunnecessary power consumption.

Next, in the example of FIG. 15, the beacon frame is missed when the STAhaving GID 1 (that is, belonging to group 1) wakes up. That is, the STAwhich does not receive the beacon frame including the GID (or PID)assigned thereto is continuously in the awake state until the beaconframe including the GID (or PID) thereof is received. That is, althoughthe STA wakes up at channel access interval assigned thereto, the STAcannot confirm whether the GID (or PID) thereof is included in the TIMtransmitted via the beacon frame and thus cannot confirm whether thetiming corresponds to the channel access interval assigned to the groupthereof.

In the example of FIG. 15, the STA which is switched from the sleepstate to the awake state is continuously in the awake state until thefourth beacon frame including the GID (that is, GID 1) thereof isreceived after the first beacon frame has been missed, thereby causingunnecessary power consumption. As a result, after unnecessary powerconsumption, the STA may receive the beacon frame including GID 1 andthen may perform RTS transmission, CTS reception, data frametransmission and ACK reception.

FIG. 16 shows the case in which an STA wakes up at a channel accessinterval for another group. For example, the STA having GID 3 may wakeup at the channel access interval for GID 1. That is, the STA having GID3 unnecessarily consumes power until the beacon frame having the GIDthereof is received after waking up. If a TIM indicating GID 3 isreceived via a third beacon frame, the STA may recognize the channelaccess interval for the group thereof and perform data transmission andACK reception after CCA through RTS and CTS.

3. First Proposed Method for CFO Estimation

As interest in future Wi-Fi and demand for improvement of throughput andQoE (quality of experience) after 802.11ac increase, it is necessary todefine a new frame format for future WLAN systems. The most importantpart in a new frame format is a preamble part because design of apreamble used for synchronization, channel tracking, channel estimation,adaptive gain control (AGC) and the like may directly affect systemperformance.

In the future Wi-Fi system in which a large number of APs and STAssimultaneously access and attempt data transmission and reception,system performance may be limited when legacy preamble design isemployed. That is, if each preamble block (e.g., a short training field(STF) in charge of AGC, CFO estimation/compensation, timing control andthe like or a long training field (LTF) in charge of channelestimation/compensation, residual CFO compensation and the like)executes only the function thereof defined in the legacy preamblestructure, frame length increases, causing overhead. Accordingly, if aspecific preamble block can support various functions in addition to thefunction designated therefor, an efficient frame structure can bedesigned.

Furthermore, since the future Wi-Fi system considers data transmissionin outdoor environments as well as indoor environments, the preamblestructure may need to be designed differently depending on environments.Although design of a unified preamble format independent of environmentvariation can aid in system implementation and operation, of course, itis desirable that preamble design be adapted to system environment.

Preamble design for efficiently supporting various functions isdescribed hereinafter. For convenience, a new WLAN system is referred toas an HE (High Efficiency) system and a frame and a PPDU (PLCP (PhysicalLayer Convergence Procedure) Protocol Data Unit) of the HE system arerespectively referred to as an HE frame and an HE PPDU. However, it isobvious to those skilled in the art that the proposed preamble isapplicable to other WLAN systems and cellular systems in addition to theHE system.

The following table 1 shows OFDM numerology which is a premise of apilot sequence transmission method described below. Table 1 shows anexample of new OFDM numerology proposed in the HE system and numeralsand items shown in Table 1 are merely examples and other values may beapplied. Table 1 is based on the assumption that FFT having a size fourtimes the legacy one is applied to a given BW and 3 DCs are used per BW.

TABLE 1 Parameter CBW20 CBW40 CBW80 CBW80 + 80 CBW160 DescriptionN_(FFT) 256 512 1024 1024 2048 FFT size N_(SD) 238 492 1002 1002 2004Number of complex data numbers per frequency segment N_(SP) 4 6 8 8 16Number of pilot values per frequency segment N_(ST) 242 498 1010 10102020 Total number of subcarriers per frequency segment. See NOTE. N_(SR)122 250 506 506 1018 Highest data subcarrier index per frequency segmentN_(Seg) 1 1 1 2 1 Number of frequency segments Δ_(F) 312.5 kHzSubcarrier frequency Spacing for non-HE portion Δ_(F) _(—) _(HE) 78.125kHz Subcarrier frequency Spacing for HE portion T_(DFT) 3.2 μs IDFT/DFTperiod for non-HE portion T_(DFT) _(—) _(HE) 12.8 μs IDFT/DFT period forHE portion T_(GI) 0.8 μs = T_(DFT)/4 Guard interval duration for non- HEportion T_(GI) _(—) _(HE) 3.2 μs = T_(DFT) _(—) _(HE)/4 Guard intervalduration for HE portion T_(GI2) 1.6 μs Double guard interval for non-HEportion T_(GIS) _(—) _(HE) 0.8 μs = T_(DFT) _(—) _(HE)/16 [Alternative:0.4 μs ( 1/32 CP)] Short guard interval Duration (used only for HE data)T_(SYML) 4 μs = T_(DFT) + T_(GI) Long GI symbol interval for non-HEportion T_(SYML) _(—) _(HE) 16 μs = T_(DFT) _(—) _(HE) + T_(GI) _(—)_(HE) Long GI symbol interval for HE portion T_(SYMS) _(—) _(HE) 13.6 μs= T_(DFT) _(—) _(HE) + T_(GIS) _(—) _(HE) [Alternative: 13.2 μs Short GIsymbol (with 1/32 CP)] interval (used only for HE data) T_(SYM) T_(SYML)or T_(SYMS) depending on the GI used Symbol interval for non-HE portionT_(SYM) _(—) _(HE) T_(SYML) _(—) _(HE) or T_(SYMS) _(—) _(HE) dependingon the GI used Symbol interval for HE portion T_(L-STF) 8 μs = 10 *T_(DFT)/4 Non-HE Short Training field duration T_(L-LTF) 8 μs = 2 ×T_(DFT) + T_(GI2) Non-HE Long Training field duration T_(L-SIG) 4 μs =T_(SYML) Non-HE SIGNAL field duration T_(HE-SIGA) 12.8 μs = 2(T_(SYML) +3T_(GI)) in HE-PPDU format-1 or HE Signal A field T_(SYML) _(—) _(HE) inHE-PPDU format-2 duration and HE-PPDU format-3 T_(HE-STF) T_(SYML) _(—)_(HE) HE Short Training field duration T_(HE-LTF) T_(SYML) _(—) _(HE)Duration of each HE LTF symbol T_(HE-SIGB) T_(SYML) _(—) _(HE) HE SignalB field duration N_(service) 16 Number of bits in the SERVICE fieldN_(tail)  6 Number of tail bits per BCC encoder NOTE N_(ST) = N_(SD) +N_(SP)

FIG. 17 is a diagram illustrating frame structures related to anembodiment of the present invention. As illustrated in FIGS. 17(a),17(b) and 17(c), various frame structures can be configured, and aproposed pilot sequence transmission method is related to an HE-STF(High Efficiency Short Training Field) in a preamble in a framestructure.

FIGS. 18 and 19 are diagrams illustrating frame structures according toanother embodiment of the present invention and constellations thereof.Specifically, FIG. 18 (a) illustrates a time-domain frame structure of ahigh throughput (HT) system based on 802.11n. In FIG. 18 (a), L-SIG andHT-SIG indicates a legacy signal field and a high throughput signalfield, respectively. Assuming that one OFDM symbol length is 4 us, theL-SIG corresponds to a single OFDM symbol and the HT-SIG corresponds totwo OFDM symbols.

Meanwhile, assuming that data transmission is performed based on theframe structure shown in FIG. 18 (a), system information can be mappedto constellations shown in FIG. 18 and then transmitted to a UE.

FIG. 19 (a) shows a frame structure of a very high throughput (VHT)system based on 802.11ac. Similar to FIG. 18, in the VHT system, systeminformation is mapped to constellations shown in FIG. 19 (b) and thentransmitted to a UE using L-SIG and VHT-SIG-A fields shown in FIG. 19(a).

FIG. 20 is a diagram illustrating frequency-domain pilot signals relatedto the proposed embodiments.

After receiving such fields as L-SIG, HT-SIG, and VHT-SIG-A from a BS(or transmission module) as described with reference to FIGS. 18 and 19,a UE (or reception module) performs Fast Fourier Transform (FFT)operation. The results of the operation can be expressed as shown inFIG. 20.

FIG. 20 shows converted frequency-domain pilot signals in each OFDMsymbol. In FIG. 20, a signal received through each subcarrier can beexpressed as shown in Equation 1.

r _(k) ^(n) =H _(k) ^(n) s _(k) ^(n)  [Equation 1]

In Equation 1, k denotes a subcarrier index and n denotes an OFDM symbolindex. In addition, H_(k) ^(n) indicates a channel between an n^(th)OFDM symbol and a k^(th) subcarrier. Assuming that a data signaltransmitted through H_(k) ^(n) is s_(k) ^(n), a received signal can beexpressed as r_(k) in Equation 1.

Meanwhile, as shown in FIG. 20, some subcarriers include guard intervalsor direct current (DC) components and such carriers are set to nullwithout loading data signals. On the other hand, in case of subcarriersin which data signals are loaded, a set of their indices is defined asC.

Before describing the proposed CFO estimation method, the concept of aCFO is explained hereinbelow. The CFO (carrier frequency offset) occursdue to performance of oscillators included in both a transmission moduleand a reception module or Doppler effects. The CFO can be divided intoan integer part and a fractional part (for example, if the CFO has avalue of 2.5, the integer part is 2 and the fractional part is 0.5). Asubcarrier is circular shifted by the integer part of the CFO, but thefractional part of the CFO causes interference between subcarriers.

In the HT and VHT systems, a reception module estimates a CFO valueusing an L-STF field and an L-LTF field. After the CFO estimation, theestimated result is applied to a received OFDM symbol. By doing so, theeffect of the CFO is eliminated as shown in Equation 2.

$\begin{matrix}\begin{matrix}{{{D\left( {- \hat{\epsilon}} \right)}y} = {{D\left( {- \hat{\epsilon}} \right)}\left( {{{D(\epsilon)}x} + n} \right)}} \\{= {{{D\left( {\epsilon - \hat{\epsilon}} \right)}x} + n^{\prime}}} \\{= {{{D\left( {\Delta \; \epsilon} \right)}x} + n^{\prime}}}\end{matrix} & \left\lbrack {{Equation}\mspace{14mu} 2} \right\rbrack\end{matrix}$

In Equation 2, ϵ indicates an actual CFO value and {circumflex over (ϵ)}indicates an estimated CFO value. In addition, y indicates a receivedsignal vector when the CFO is present, x indicates a received signalvector when the CFO is not present, and n indicates a noise vector. Adiagonal matrix D of Equation 2 is defined as shown in Equation 3.

$\begin{matrix}{{D(\epsilon)} = \begin{bmatrix}1 & 0 & \; & 0 \\0 & e^{j\; 2\; \pi \; {\epsilon/N}} & \ddots & \; \\\; & \ddots & \ddots & 0 \\0 & \; & 0 & e^{j\; 2\; \pi \; {{\epsilon {({N - 1})}}/N}}\end{bmatrix}} & \left\lbrack {{Equation}\mspace{14mu} 3} \right\rbrack\end{matrix}$

If the reception module perfectly estimates the CFO value using theL-SFT and the L-LTF (ϵ={circumflex over (ϵ)}), the reception module canperfectly eliminate the CFO from the received signal using Equation 2and Equation 3 (Δϵ=0). However, since the CFO is slightly changeddepending on time, it is difficult to perfectly estimate the CFO value(ϵ≠{circumflex over (ϵ)}). Thus, a residual CFO can be defined as shownin Equation 4.

Δϵ=ϵ−{circumflex over (ϵ)}+{tilde over (ϵ)}  [Equation 4]

In Equation 4, {tilde over (ϵ)} indicates a CFO value changed dependingon time. To re-estimate the residual CFO, the reception module utilizespilot signals included in the L-SIG and HT-SIG. In the HT system, theresidual CFO is estimated using four pilot signals. However, since theHT system has a relatively small number of pilot signals, performance ofthe CFO estimation is significantly decreased in case of a low SNR. Thatis, the number of pilot signal needs to be increased to overcome such aproblem but it may cause throughput reduction as a trade-off. Therefore,a CFO estimation method for minimizing performance degradation in caseof a low SNR while maintaining an HT system structure needs to bedeveloped.

Hereinafter, a CFO method according to the present invention will bedescribed with reference to FIGS. 21 to 23. According to proposedembodiments, the reception module can estimate the CFO in a blindmanner, i.e., using a data signal instead of a pilot signal.

FIGS. 21 and 22 are diagrams for explaining a proposed CFO estimationmethod. In FIGS. 21 and 22, it is assumed that data transmission isperformed according to a binary phase shift keying (BPSK) scheme or aquadrature phase shift keying (QBPSK) scheme.

According to the proposed CFO estimation method, y_(k) ^(n), whichreflects a received signal in two consecutive OFDM symbols based onEquation 1, can be defined as shown in Equation 5.

y _(k) ^(n) ≙r _(k) ^(n+1)(r _(k) ^(n))*, k∈C, n=1,2, . . .,L  [Equation 5]

In Equation 5, L is defined as (the number of total OFDM symbols towhich the proposed CFO estimation method is applied—1). For example,when two OFDM symbols are used as shown in FIG. 20, L is set to 1. Whenthree OFDM symbols are used for the L-SIG and the HT-SIG shown in FIG.18 (a), L is set to 3.

Hereinafter, Equation 5 is described in detail. In the case of twoconsecutive OFDM symbols (i.e., n^(th) OFDM symbol and (n+1)^(th) OFDMsymbol), a channel is not rapidly changed. In other words, in Equation5, y_(k) ^(n) is defined on the assumption that the two consecutive OFDMsymbols have the same channel.

According to the proposed CFO estimation method, a process shown inEquation 6 below is performed after calculation of y_(k) ^(n).

$\begin{matrix}{z_{k}^{n} = \left\{ \begin{matrix}y_{k}^{n} & {{{if}\mspace{14mu} {real}\mspace{11mu} \left( y_{k}^{n} \right)} \geq 0} \\{- y_{k}^{n}} & {{{if}\mspace{14mu} {real}\mspace{11mu} \left( y_{k}^{n} \right)} < 0}\end{matrix} \right.} & \left\lbrack {{Equation}\mspace{14mu} 6} \right\rbrack\end{matrix}$

In Equation 6, z_(k) ^(n) is determined by a sign of a real part ofy_(k) ^(n). In addition, Equation 7 below shows a process fordetermining a final residual CFO.

$\begin{matrix}{{\Delta \; \hat{\epsilon}} = {{{angle}\left( {\sum\limits_{n = 1}^{L}{\sum\limits_{k \in C}z_{k}^{n}}} \right)} \times \frac{N}{2\; {\pi \left( {N + N_{g}} \right)}}}} & \left\lbrack {{Equation}\mspace{14mu} 7} \right\rbrack\end{matrix}$

In equation 7, Δ{circumflex over (ϵ)} indicates a finally calculatedresidual CFO value, and N and N_(g) indicate an OFDM symbol length and acyclic prefix (CP) length, respectively. It can be seen from Equation 7that the processes described in Equation 5 and Equation 6 are performedwith respect to the entirety of the set C consisting of subcarrierswhere data are loaded.

Hereinafter, processes for Equations 5 to 7 are described in detail.Assuming that n is equal to 1 in Equation 5, y_(k) ¹ can be expressed asshown in Equation 8.

$\begin{matrix}\begin{matrix}{y_{k}^{1}\overset{\Delta}{=}{r_{k}^{2}\left( r_{k}^{1} \right)}^{*}} \\{\approx {H_{k}^{2}{s_{k}^{2}\left( {H_{k}^{1}s_{k}^{1}} \right)}^{*}e^{j\; 2\; \pi \; \Delta \; {{\epsilon {({N + N_{g}})}}/N}}}} \\{= {{H_{k}^{2}\left( H_{k}^{1} \right)}^{*}{s_{k}^{2}\left( s_{k}^{1} \right)}^{*}e^{j\; 2\; \pi \; \Delta \; {{\epsilon {({N + N_{g}})}}/N}}}} \\{\approx {\rho {H_{k}^{1}}^{2}{{sign}\left( {s_{k}^{2}\left( s_{k}^{1} \right)}^{*} \right)}e^{j\; 2\; \pi \; \Delta \; {{\epsilon {({N + N_{g}})}}/N}}}}\end{matrix} & \left\lbrack {{Equation}\mspace{14mu} 8} \right\rbrack\end{matrix}$

In Equation 8, ρ indicates a power component of the data signal s_(k)^(n). In addition, a function sign(a) has a value of 1 when a variable ahas a positive sign and a value of −1 when the variable a has a negativesign. Meanwhile, approximation in the second line of Equation 8 isachieved based on the assumption that interference between subcarrierscaused by the residual CFO can be ignored. Moreover, approximation inthe fourth line is achieved on the assumption that channels H_(k) ¹ andH_(k) ² in the two OFDM symbols are equal to each other. Consideringthat the BPSK scheme is applied together with the above assumptions, anequation of s_(k) ²(s_(k) ¹)*=ρsign(s_(k) ²(s_(k) ¹)*) is alwayssatisfied.

In Equation 8, when an equation of ρsign(s_(k) ²(s_(k) ¹)*)=ρ issatisfied, y_(k) ¹ is defined as shown in Equation 9.

y _(k) ¹ ≈ρ|H _(k) ¹|² e ^(j2πΔϵ(N+N) ^(g) ^()/N)  [Equation 9]

Referring to FIG. 21, it can be seen that a component ρ|H_(k) ¹|²corresponds to a radius of the illustrated circle and a component2πΔϵ(N+N_(g))/N corresponds to a phase value of the illustrated point.In this case, a phase value of y_(k) ¹ is a function of the residual CFO(Δ{circumflex over (ϵ)}) and the value is proportional to the residualCFO value. For example, if the residual CFO value is 0, the phase ofy_(k) ¹ is also 0. If the phase of y_(k) ¹ is smaller than 2π, a ratioof the residual CFO to the phase of y_(k) ¹ is 1:1. Thus, it is possibleto estimate the residual CFO value from the phase of y_(k) ¹.

On the other hand, when an equation of is ρsign(s_(k) ²(s_(k) ¹)*)=−ρ issatisfied, Equation 9 is expressed as shown in Equation 22 instead ofEquation 21. If the reception module is aware of ρsign(s_(k) ²(s_(k)¹)*)=−ρ, the reception module can estimate the residual CFO from thephase of y_(k) ¹ through a simple calculation. However, such acalculation is disadvantageous in that the reception module should knowreceived data before performing the calculation. Moreover, according tothe proposed CFO method, since the CFO estimation is performed in ablind manner without the use of a pilot signal, whether a value ofsign(s_(k) ²(s_(k) ¹)*) is positive or negative should be firstdetermined to accurately estimate the residual CFO.

To solve the above-mentioned problem, a case where the residual CFOvalue is relatively small compared to the total CFO is considered. Inother words, first of all, the reception module may estimate the CFOvalue using a preamble part such as the L-STF and the L-LTF and thenestimate the residual CFO value based on the L-SIG and the HT-SIG. Whenthe CFO value is approximately estimated through the primary CFOestimation process as described above, a phase of the residual CFO has arelatively small value and thus, a range of the phase of y_(k) ¹ alsodoes not have a large value. Accordingly, Equation 10 can be deduced asfollows.

sign(s _(k) ²(s _(k) ¹)*)=1 if angle(y _(k) ¹)ϵ{first quadrant,fourthquadrant}

sign(s _(k) ²(s _(k) ¹)*)=−1 if angle(y _(k) ¹)ϵ{second quadrant,thirdquadrant}  [Equation 10]

If the phase of y_(k) ¹ belongs to the first quadrant or the fourthquadrant, the equation of ρsign(s_(k) ²(s_(k) ¹)*)=ρ is satisfied. Onthe contrary, if the phase of y_(k) ¹ belongs to the second quadrant orthe third quadrant, the equation of ρsign(s_(k) ²(s_(k) ¹)*)=−ρ issatisfied. That is, Equation 6 can be explained by Equation 10, i.e., aresidual CFO relationship. In other words, according to Equation 6, inthe case of ρsign(s_(k) ²(s_(k) ¹)*)=−ρ, the phase of y_(k) ¹ is changedby π.

Meanwhile, when there is no noise, all phases of z_(k) ¹, k∈C, which arecalculated according to Equation 6, have values of 2πΔϵ(N+N_(g))/N. Thatis, calculation results of all subcarriers are in a state of in-phase(or co-phase). According to Equation 7, the residual CFO can beaccurately estimated when there is noise. In Equation 7, the phase andpower of the signal are added all together, a robust result with respectto a dominant noise can be obtained.

According to the above-described method, the total CFO can be accuratelymeasured by estimating the residual CFO in a blind manner. In addition,even when the SNR/SINR is low, overhead due to transmission of anadditional pilot signal does not occur and thus, communication can beefficiently performed.

Meanwhile, according to an embodiment of the present invention, the CFOmethod can be applied to a subset C′ of the subcarrier set C where datais loaded. That is, although it can be seen in Equation 7 that theprocess for estimating the residual CFO is performed by adding phases ofall samples, the residual CFO can be estimated using only somesubcarriers instead of the total subcarrier samples. Accordingly, asubset of the set C where data signals are loaded is defined as thesubset C′ and then the set C can be substituted with the subset C′ inEquation 7.

When a specific subcarrier is significantly faded, a size of datareceived through the corresponding subcarrier is also significantlydecreased. In this case, compared to other data samples, thecorresponding data sample is rarely attributed to the residual CFOestimation. In other words, the residual CFO can be estimated using onlyrelatively large sizes of data samples. In addition, even if small sizesof samples are excluded, it does not cause performance degradation.

According to an embodiment generated by modifying the above-describedembodiment, if the reception module knows sizes of data signals receivedthrough respective subcarriers, the reception module can arrange thesubcarriers in descending order of sizes and then define subcarrierswith sizes greater than a threshold as the subset C′. According to thisembodiment, since the reception module may skip the processes describedin Equations 5 to 7 (i.e., complexity associated with the processesdescribed in Equations 5 to 7 may be reduced), performance degradationin the residual CFO estimation can also be minimized.

The present invention has been described with reference to cases whereeither the BPSK scheme or the QBPSK scheme is applied. However, theinvention can be applied to a case where the BPSK scheme and the QBPSKscheme are alternately applied to each OFDM symbol. However, when twoconsecutive OFDM symbols are mapped to different constellations, theproposed method cannot be applied as it is because a product of twosignals is not 1 or −1.

In this case, the residual CFO can be estimated according to Equation11.

$\begin{matrix}{{y_{k}^{n}\overset{\Delta}{=}{{\overset{\sim}{r}}_{k}^{n + 1}\left( {\overset{\sim}{r}}_{k}^{n} \right)}^{*}},{k \in C},{n = 1},2,\ldots \mspace{14mu},L} & \left\lbrack {{Equation}\mspace{14mu} 11} \right\rbrack \\{{\overset{\sim}{r}}_{k} = \left\{ \begin{matrix}{r_{k}e^{{- j}\; {\pi/2}}} & {{if}\mspace{14mu} s_{k}\mspace{14mu} {is}\mspace{14mu} {QBPSK}} \\r_{k} & {otherwise}\end{matrix} \right.} & \;\end{matrix}$

Equation 5 can be substituted with Equation 11. In Equation 11, a phaseof a signal received in the OFDM symbol where the QBPSK scheme is usedis shifted by e^(−jπ/2) but a phase of a signal received in the OFDMsymbol where the BPSK scheme is used is not shifted. According toEquation 11, when the BPSK and QBPSK are alternately used in a series ofOFDM symbols, it is possible to obtain the same results as those inEquations 6 and 7.

FIG. 23 is a flowchart for explaining the proposed CFO estimationmethod. Specifically, FIG. 23 shows a time series flow of the CFOestimation method according to the aforementioned embodiments. Thus, itis apparent that although the aforementioned features described withreference to FIGS. 18 to 22 are not explicitly shown and described inFIG. 23, the features can be applied to the flowchart in FIG. 23 in thesame or similar manner.

First, a transmission module transmits data to a reception module 52310.In this case, data is transmitted in a unit of frame, which is definedby an OFDM symbol and a subcarrier. In addition, the data is mapped to aspecific constellation and then transmitted to the reception module. Assuch as a constellation, either BPSK or QBPSK may be used.Alternatively, the BPSK and QBPSK may be alternately used in a series ofconsecutive OFDM symbols.

Meanwhile, the reception module primarily estimate a CFO value based ona received signal [S2320]. Such a process is performed using a preamblepart such as an L-STF, an L-LTF, and the like. However, since a CFO ischanged depending on time, the CFO value estimated in the step S2320 maybe inaccurate.

Therefore, the reception module estimate a residual CFO value tocompensate the primarily estimated CFO value [S2330]. As describedabove, the reception module estimates the residual CFO value on theassumption that channels of received signals in two consecutive OFDMsymbols are equal to each other. Specifically, the reception modulecalculate a product of the two received signal and then checks a sign ofa real part of the product based on the assumption that the residual CFOvalue is smaller than the primarily estimated CFO value. If totalsubcarriers are in a state of in-phase, the reception module can obtainthe residual CFO from a phase value calculated by adding results of allsubcarriers.

Finally, the reception module can accurately decode the data transmittedfrom the transmission module by eliminating effects of the CFO estimatedin the step S2320 and the residual CFO estimated in the step S2330 fromthe received signal.

4. Second Proposed Method for CFO Estimation

FIGS. 24 to 26 are diagrams illustrating a CFO estimation methodaccording to a proposed embodiment. That is, the CFO estimation methodwhen the BPSK and/or QBPSK scheme is used has been described above.Next, a description will be given of a CFO estimation method when datais transmitted using a quadrature phase shift keying (QPSK) scheme.

Similar to the process described in Equation 5, y_(k) ^(n), whichreflects signals received in two consecutive OFDM symbols based onEquation 1, can be defined as shown in Equation 12.

y _(k) ^(n) ≙r _(k) ^(n+1)(r _(k) ^(n))*, k∈C, n=1,2, . . .,L  [Equation 12]

In Equation 12, L is defined as (the number of total OFDM symbols towhich the proposed CFO estimation method is applied −1). For example,when two OFDM symbols are used as shown in FIG. 20, L is set to 1 (L=1).When three OFDM symbols are used similar to the L-SIG and the HT-SIG ofFIG. 18 (a), L is set to 3 (L=3).

Hereinafter, Equation 12 is described in detail. In the case of twoconsecutive OFDM symbols (i.e., n^(th) OFDM symbol and (n+1)^(th) OFDMsymbol), a channel is not rapidly changed. In other words, in Equation12, y_(k) ^(n) is defined on the assumption that the two consecutiveOFDM symbols have the same channel.

Thereafter, if y_(k) ^(n) is calculated at a reception module, acalculation can be performed as shown in Equation 12 below.

$\begin{matrix}{z_{k}^{n} = \left\{ \begin{matrix}y_{k}^{n} & {{{{if}\mspace{14mu} {{real}\left( y_{k}^{n} \right)}} \geq 0},{{{real}\left( y_{k}^{n} \right)} \geq {{imag}\left( y_{k}^{n} \right)}}} \\{- y_{k}^{n}} & {{{{if}\mspace{14mu} {real}\; \left( y_{k}^{n} \right)} < 0},{{{real}\left( y_{k}^{n} \right)} \geq {{imag}\left( y_{k}^{n} \right)}}} \\{{- j} \times y_{k}^{n}} & {{{{if}\mspace{14mu} {{imag}\left( y_{k}^{n} \right)}} \geq 0},{{{real}\left( y_{k}^{n} \right)} < {{imag}\left( y_{k}^{n} \right)}}} \\{j \times y_{k}^{n}} & {{{{if}\mspace{14mu} {{imag}\left( y_{k}^{n} \right)}} < 0},{{{real}\left( y_{k}^{n} \right)} < {{imag}\left( y_{k}^{n} \right)}}}\end{matrix} \right.} & \left\lbrack {{Equation}\mspace{14mu} 13} \right\rbrack\end{matrix}$

In equation 13, z_(k) ^(n) is determined by signs and magnitudes of areal part and an imaginary part of y_(k) ^(n). Although details will bedescribed later, the four different cases in Equation 13 mayrespectively correspond to four different quadrants of theconstellation. Thus, the function z_(k) ^(n) can be generated indifferent ways as shown in Equation 13 depending on a quadrant to whichy_(k) ^(n) belongs.

Next, Equation 14 shows a process for determining a final residual CFObased on Equation 13.

$\begin{matrix}{{\Delta \; \hat{\epsilon}} = {{angle}\; \left( {\sum\limits_{n = 1}^{L}{\sum\limits_{k \in C}z_{k}^{n}}} \right) \times \frac{N}{2\; {\pi \left( {N + N_{g}} \right)}}}} & \left\lbrack {{Equation}\mspace{14mu} 14} \right\rbrack\end{matrix}$

In Equation 14, Δ{circumflex over (ϵ)} indicates a final residual CFOvalue calculated at a reception module, and N and N_(g) indicate an OFDMsymbol length and a cyclic prefix (CP) length, respectively. It can beseen from Equation 14 that the processes described in Equation 12 andEquation 13 are performed with respect to the entirety of the set Cconsisting of subcarriers where data are loaded.

Hereinafter, details of the processes in Equations 12 to 14 aredescribed. First, y_(k) ¹ can be expressed as shown in Equation 15according to the approximation procedure mentioned in Equation 8

y _(k) ¹ ≈|H _(k) ¹|² s _(k) ²(s _(k) ¹)*e ^(j2πΔϵ(N+N) ^(g)^()/N)  [Equation 15]

When data transmission is performed using the QPSK scheme, s_(k) ²(s_(k)¹)* can be distributed to one of the four points shown in FIG. 24. Inother words, a phase of the product of data transmitted in the twoconsecutive OFDM symbols may be one of the four points of theconstellation shown in FIG. 24. Meanwhile, in FIG. 24, the value ofs_(k) ²(s_(k) ¹)* is defined as g_(k) ¹ according to Equation 16.

g _(k) ¹ =s _(k) ²(s _(k) ¹)*[Equation 16]

It can be seen from FIG. 24 that a phase of g_(k) ¹ is one of {0, π/2,π, π/3}.

Meanwhile, if a residual CFO value to be measured at the receptionmodule is relatively smaller than a total CFO value, y_(k) ¹ may havevalues shown in FIG. 25. That is, the residual CFO has a relativelysmaller value than a CFO value firstly estimated based on the preamblepart. Thus, it can be known that the phase of the function y_(k) ¹ usedfor residual CFO estimation does not have a relatively large value.Further, the phase of y_(k) ¹ may have values that are not significantlydifferent from phase values at the four points of the QPSKconstellation.

If the assumption that the residual CFO has a relatively small value isnot established, a phase ambiguity problem may occur while the residualCFO is measured. However, based on the aforementioned assumption, it canbe seen that the phase change of y_(k) ¹ due to the residual CFO iswithin the range of {0, π/2} (i.e., the first embodiment of FIG. 25),{π/2, π} (i.e., the second embodiment of FIG. 25), {π, 3π/2} (the thirdembodiment of FIG. 25), or {3π/2, 2π} (the fourth embodiment of FIG.25).

Equation 17 can be deduced based on the above results.

$\begin{matrix}\left. y_{k}^{1}\Rightarrow\left\{ \begin{matrix}{◯1} & {{{{if}\mspace{14mu} {{real}\left( y_{k}^{1} \right)}} \geq 0},{{{real}\left( y_{k}^{1} \right)} \geq {{imag}\left( y_{k}^{1} \right)}}} \\{◯3} & {{{{if}\mspace{14mu} {real}\; \left( y_{k}^{1} \right)} < 0},{{{real}\left( y_{k}^{1} \right)} \geq {{imag}\left( y_{k}^{1} \right)}}} \\{◯2} & {{{{if}\mspace{14mu} {{imag}\left( y_{k}^{1} \right)}} \geq 0},{{{real}\left( y_{k}^{1} \right)} < {{imag}\left( y_{k}^{1} \right)}}} \\{◯4} & {{{{if}\mspace{14mu} {{imag}\left( y_{k}^{1} \right)}} < 0},{{{real}\left( y_{k}^{1} \right)} < {{imag}\left( y_{k}^{1} \right)}}}\end{matrix} \right. \right. & \left\lbrack {{Equation}\mspace{14mu} 17} \right\rbrack\end{matrix}$

That is, it is possible to obtain a quadrant to which y_(k) ¹ belongsamong the four quadrants of the constellation by comparing/analyzing areal part and an imaginary part of y_(k) ¹ according to Equation 17. Forexample, when y_(k) ¹ satisfies the condition of real(y_(k) ¹)≥0,real(y_(k) ¹)≥imag(y_(k) ¹), y_(k) ¹ corresponds to the first embodimentof FIGS. 17 and 25 (i.e., g_(k) ¹=ρ). Consequently, when the receptionmodule fails to decode data of y_(k) ¹ correctly, it is possible toestimate the phase of g_(k) ¹ from the phase of y_(k) ¹. In other words,in this case, even though the CFO is estimated in a blind decodingmanner, the phase ambiguity problem does not occur.

Meanwhile, after a case to which y_(k) ¹ belongs among the four cases isconfirmed, the function z_(k) ^(n) can be generated through processingof y_(k) ^(n) as shown in Equation 13. Since all values of z_(k) ^(n) isin a state of in-phase (or co-phase), the final residual CFO can beestimated based on the generated z_(k) ^(n) and FIG. 14.

Meanwhile, according to the proposed embodiment, as |H_(k) ¹|² decreasesand the residual CFO decreases, the accuracy of the residual CFOestimation can be improved because noise effects can be eliminated fromthe final residual CFO.

According to another proposed embodiment, the BPSK and the QPSK can bealternately used in a series of OFDM symbols. That is, similar to thecase in which the BPSK scheme and the QBPSK scheme are alternately used,the BPSK and the QPSK can be alternately used in two OFDM symbols.However, in this case, since the phase of the product of two receivedsignals is not placed at one of the four points shown in FIG. 24, theaforementioned method cannot be applied as it is.

In this case, the residual CFO can be estimated according to Equation18.

$\begin{matrix}{{y_{k}^{n}\overset{\Delta}{=}{{\overset{\sim}{r}}_{k}^{n + 1}\left( {\overset{\sim}{r}}_{k}^{n} \right)}^{*}},{k \in C},{n = 1},2,\ldots \mspace{14mu},L} & \left\lbrack {{Equation}\mspace{14mu} 18} \right\rbrack \\{{\overset{\sim}{r}}_{k}^{l} = \left\{ \begin{matrix}{r_{k}^{l}e^{j\frac{\pi}{4}}} & {{if}\mspace{14mu} B\; P\; S\; K} \\r_{k}^{l} & {otherwise}\end{matrix} \right.} & \;\end{matrix}$

Equation 18 can be used instead of Equation 12. According to Equation18, a phase of a signal received in the OFDM symbol where the BPSK isused can be uniformly changed by π/4. In this case, as shown in FIG. 26,a constellation of the BPSK of which the phase is changed partiallymatches that of the QPSK. Thus, the aforementioned embodiments can beequally applied to processes after Equation 18.

Meanwhile, in this embodiment, an angle for the phase change can bedefined as

${\alpha \frac{\pi}{4}},\left( {{\alpha = {{2n} + 1}},{n \in {\mathbb{Z}}}} \right)$

(where Z is a set of integers). That is, it is meaningful that the BPSKconstellation is changed as a part of the QPSK constellation and achanged phase value may be different. Moreover, according to theaforementioned embodiments, phase values of r_(k) ^(l) in all OFDMsymbols where the BPSK is used can be uniformly changed by

${\alpha \frac{\pi}{4}},{\left( {{\alpha = {{2n} + 1}},{n \in {\mathbb{Z}}}} \right).}$

However, even when the phases of are r_(k) ^(l) rotated using differentvalues, the result may be the same as the above result. For instance,when a phase value of r₁ ^(l) is rotated by π/4 and a phase value of r₂^(l) is rotated by 3π/4, the result may be the same as that of the casein which two received signals are rotated by the same phase valueaccording to Equation 18. This is because points of the BPSKconstellation are simply moved to points of the QPSK constellation.

The aforementioned embodiment can be equally applied to a case where theQBPSK and the QPSK are alternately used in each OFDM symbol. That is,when the QBPSK and the QPSK are used in two consecutive OFDM symbols,the aforementioned CFO estimation procedure can be equally applied bychanging a phase value of a symbol where the QBPSK is used instead ofthe BPSK.

According to a further embodiment, when the BPSK and QPSK is used in aseries of OFDM symbols, Equation 19 can be used instead of Equation 12.

$\begin{matrix}{{y_{k}^{n}\overset{\Delta}{=}{{r_{k}^{n + 1}\left( r_{k}^{n} \right)}^{*}e^{j\frac{\pi}{4}}}},{k \in C},{n = 1},2,\ldots \mspace{14mu},L} & \left\lbrack {{Equation}\mspace{14mu} 19} \right\rbrack\end{matrix}$

Unlike Equation 18, according to Equation 19, the entire phase of y_(k)^(n) is changed by π/4 instead of changing phases of individual OFDMsymbols. Equation 19 can be satisfied irrespective of whether the BPSKand the QPSK is used for an n^(th) OFDM symbol. This is because the BPSKand the QPSK are alternately used in every two consecutive OFDM symbols.Thus, it is possible to obtain the same result as that of Equation 18.

In Equation 19, even when phases of y_(k) ^(n) are rotated by differentvalues, the same result occurs. For example, when a phase of y_(l) ^(n)is rotated by π/4 and a phase of y₂ ^(n) is rotated by 3π/4, the sameresult occurs. In addition, the embodiment described with reference toEquation 19 can be equally applied to not only the case where the BPSKand the QPSK are alternately used in every two OFDM symbols but also thecase where the QBPSK and the QPSK are alternately used in every two OFDMsymbols.

Hereinafter, a description will be given of UE's operation related tothe aforementioned embodiments. First, while data transmitted from atransmission module is received at a reception module, a CFO occurringin an n^(th) OFDM symbol is defined as ϵ_(n). To eliminate the CFO inthe n^(th) time-domain OFDM symbol and a CFO in an (n+1)^(th)time-domain OFDM symbol, the reception module estimates CFO(s) using apreamble part of a firstly received frame. The estimated CFO is definedas {circumflex over (ϵ)}_(n). However, since the primarily estimated CFOis not complete (i.e., ϵ_(n)≠{circumflex over (ϵ)}_(n)), there must be aresidual CFO (Δ{circumflex over (ϵ)}_(n)) Thus, the reception module canestimate the residual CFO using the aforementioned embodimentsindependently or any combination thereof.

After estimating the residual CFO, the reception module can correct aphase of a signal r_(k) ^(l) received in a subcarrier k as shown inEquation 20 to eliminate the effect of the estimated residual CFO fromthe received signal.

{tilde over (r)} _(k) ^(l) =r _(k) ^(l) e ^(−j2πΔ{tilde over (ϵ)}(N+N)^(g) ^()/N) , l=n,n+1  [Equation 20]

By doing so, the reception module can compensate phase distortion in thereceived signal due to the residual CFO and thus, reception SINR canfinally be improved.

Meanwhile, unlike a method of eliminating a CFO in the time domain,according to the method of eliminating a CFO in the frequency domain, itis impossible to cancel the effect of a leak signal which occurs due tothe CFO. Therefore, to eliminate the CFO effects from (n+2)^(th) and(n+3)^(th) time-domain OFDM symbols, a CFO estimation value shown inEquation 21 can be used.

{circumflex over (ϵ)}_(n+2)={circumflex over (ϵ)}_(n)+Δ{circumflex over(ϵ)}_(n)  [Equation 21]

In Equation 21, since {circumflex over (ϵ)}_(n+2) is closer to ϵ_(n+2)than {circumflex over (ϵ)}_(n), there may be a smaller residual CFO.Thus, a frequency-domain received signal may have less signal leakageafter FFT operation and thus efficiency can be improved in terms of areception SNR. Next, Δ{circumflex over (ϵ)}_(n+2) is estimated using theproposed embodiments and then the phase of the signal received in thesubcarrier can be corrected as described in Equation 20.

FIG. 27 is a flowchart illustrating the CFO estimation method accordingto the proposed embodiment. Specifically, FIG. 27 shows a time seriesflow of the CFO estimation method according to the embodiments describedwith reference to FIGS. 24 to 26. Thus, it is apparent that although theaforementioned features described with reference to FIGS. 18 to 22 arenot explicitly shown and described in FIG. 23, the features can beapplied to the flowchart in FIG. 23 in the same or similar manner Thus,it is apparent that although the aforementioned features described withreference to FIGS. 24 to 26 are not explicitly shown and described inFIG. 27, the features can be applied in the same or similar manner.

First, a transmission module transmits data to a reception module S2710.In this case, data is transmitted in a unit of frame, which is definedby an OFDM symbol and a subcarrier. In addition, the data is mapped to aspecific constellation and then transmitted to the reception module. Assuch as a constellation, the QPSK can be used. Moreover, the BPSK (orQBPSK) and the QPSK can be alternately used in a series of consecutiveOFDM symbols.

Meanwhile, the reception module primarily estimate a CFO value based ona received signal [S2720]. Such a process is performed using a preamblepart such as an L-STF, an L-LTF, and the like. However, since a CFO ischanged depending on time, the CFO value estimated in the step S2720 maybe inaccurate.

Therefore, the reception module estimates a residual CFO value tocompensate the primarily estimated CFO value [S2730]. As describedabove, the reception module estimates the residual CFO value on theassumption that channels of received signals in two consecutive OFDMsymbols are equal to each other. Specifically, the reception modulecalculate a product of the two received signal and then checks signs andmagnitudes of a real part and an imaginary part of the product based onthe assumption that the residual CFO value is smaller than the primarilyestimated CFO value. If signals received in the total subcarriers are ina state of in-phase, the reception module can obtain the residual CFOfrom a phase value calculated by adding results of all the subcarriers.

Finally, the reception module can accurately decode the data transmittedfrom the transmission module by eliminating effects of the CFO estimatedin the step S2720 and the residual CFO estimated in the step S2730 fromthe received signal.

5. Third Proposed Method for CFO Estimation

In the foregoing description, a method for a reception module toestimate a CFO has been explained when data transmission is performedusing BPSK (QBPSK) or QPSK. In this case, the reception moduleefficiently operates when a residual CFO has a size relatively smallerthan a size of a firstly estimated CFO. However, if the residual CFO isestimated using a blind type, the reception module is unable toguarantee that the size of the residual CFO is sufficiently small.Hence, when the size of the residual CFO is large, a method for thereception module to estimate a CFO is explained in the following.

In FIGS. 28 and 29 described in the following, a method for a receptionmodule to estimate a CFO is explained by 4 steps. FIG. 28 is a diagramillustrating a resource block for explaining a proposed embodiment. FIG.28 shows a resource block (RB) consisting of a plurality of resourceelements (REs) defined in LTE (long term evolution)/LTE-A(LTE-advanced). Referring to FIG. 28, one RB is defined by 14 OFDMsymbols arranged on a horizontal axis and 12 subcarriers arranged on avertical axis and the RB includes 168 REs in total.

In FIG. 28, REs expressed by slashes correspond to RSs (referencesignals). In this case, REs expressed by slashes slashed from the topleft to the bottom right correspond to CRSs (cell-specific RSs) and REsexpressed by slashes slashed from the top right to the bottom leftcorrespond to DMRSs (demodulation RSs). Values of the RSs are known to areception module in advance and the reception module determines CSI(channel state information) or performs channel estimation forperforming demodulation using an RS.

According to a proposed embodiment, a reception module estimates a firstresidual CFO using an RS (hereinafter, a first residual CFO). Thereception module compensates for a received data using the estimatedfirst residual CFO. In particular, when the reception module is unableto know a size of the total residual CFO, the reception module controlsa size of the residual CFO to be included within a small range.Subsequently, the reception module estimates the residual CFO accordingto the aforementioned first CFO estimation method or the second CFOestimation method (hereinafter, a second residual CFO). In other word,in order for the reception module to efficiently apply the blind typeCFO estimation method, it is necessary to control the residual CFO witha small size. Hence, the first residual CFO is estimated usingembodiments described in the following. The total residual CFO(hereinafter, a third residual CFO) corresponds to the sum of the firstresidual CFO and the second residual CFO. In the following, theembodiments proposed in the present invention are explained in detail.

First of all, a reception module estimates a first residual CFO using anRS. Similar to the aforementioned first CFO estimation method and thesecond CFO estimation method, the reception module generates a functiony_(k) ^(n) of equation 22 using signals received on two consecutive OFDMsymbols.

y _(k) ^(n) ≙r _(k) ^(n+1)(r _(k) ^(n))*, k∈D _(n) , n=1,2, . . .,L  [Equation 22]

A unique point of equation 22 different from equations 5 and 12 is inthat k corresponds to an element of D_(n) rather than an element ofC_(n). The D_(n) corresponds to a set indicating an index of asubcarrier at which an RS is positioned in an n^(th) OFDM symbol. Inparticular, the reception module generates the y_(k) ^(n) via theequation 22 using a relation between an RE at which the RS is locatedand consecutive OFDM symbols. Meanwhile, the C_(n) corresponds to a setindicating an index of a subcarrier at which data rather than the RS islocated. Since either an RS or data is transmitted only in a manner ofbeing loaded on a single RE, it satisfies {D∩C}={ϕ}.

In addition to the equation 22, equation 23 in the following describes aprocedure that the reception module measures the first residual CFO.

$\begin{matrix}{{\Delta \; {\hat{\epsilon}}_{n}^{(1)}} = {{{angle}\left( {\sum\limits_{n = 1}^{L}{\sum\limits_{k \in D_{n}}y_{k}^{n}}} \right)} \times \frac{N}{2\; {\pi \left( {N + N_{g}} \right)}}}} & \left\lbrack {{Equation}\mspace{14mu} 23} \right\rbrack\end{matrix}$

In equation 23, Δ{circumflex over (ϵ)}_(n) ⁽¹⁾ corresponds to a firstresidual CFO measured for an n^(th) OFDM symbol using an RS. Inparticular, the first residual CFO corresponds to a CFO measured foronly a subcarrier D_(n) at which an RS is located.

If the first residual CFO is determined by the equations 22 and 23, thereception module compensates for data received using the first residualCFO. The abovementioned procedure can also be comprehended as aprocedure of changing phases of the data as much as the calculated firstresidual CFO. Since the RS corresponds to a value known to the receptionmodule in advance, the first residual CFO calculated via the equations22 and 23 can compensate for the total CFO having a large value with asufficiently small value. Although the first residual CFO is notcompletely identical to the total CFO, the remaining residual CFO can bereduced to a sufficiently small value via the procedure of compensatingfor data using the first residual CFO.

Meanwhile, if the equation 23 is explained in detail, the y_(k) ^(n) canbe represented as equation 24 described in the following.

$\begin{matrix}\begin{matrix}{y_{k}^{n} \approx {{H_{k}^{n}}^{2}{s_{k}^{n + 1}\left( s_{k}^{n} \right)}^{*}e^{j\; 2\; \pi \; \Delta \; {{\epsilon_{n}{({N + N_{g}})}}/N}}}} \\\left. \rightarrow {{H_{k}^{n}}^{2}\rho \; e^{j\; 2\; \pi \; \Delta \; {{\epsilon_{n}{({N + N_{g}})}}/N}}} \right.\end{matrix} & \left\lbrack {{Equation}\mspace{14mu} 24} \right\rbrack\end{matrix}$

A first part of the equation 24 is derived from the equation 8 and theequation 9. Meanwhile, as mentioned in the foregoing description, sincean RS corresponds to a value known to the reception module in advance,the reception module knows a phase of s_(k) ^(n+1)(s_(k) ^(n))* inadvance. Hence, if a component of the s_(k) ^(n+1)(s_(k) ^(n))* iseliminated from the equation 24, a second part of the equation 24 isobtained. Since information on a phase value of the s_(k) ^(n+1)(s_(k)^(n))* component is known to the reception module in advance, it is notnecessary to have a procedure for solving a phase ambiguity problem inthe equations 22 and 23.

Meanwhile, as mentioned in the foregoing description, if the firstresidual CFO is determined, a procedure of compensating for data isrepresented as equation 25 in the following.

y _(k) ^(n) ≙r _(k) ^(n+1)(r _(k) ^(n))*e ^(−j2πΔ{tilde over (ϵ)}) ^(n)⁽¹⁾ ^((N+N) ^(g) ^()/N) , k∈D _(n) , n=1,2, . . . ,L  [Equation 25]

The equation 25 corresponds to a procedure for changing a phase of areception signal as much as the first residual CFO (Δ{circumflex over(ϵ)}_(n) ⁽¹⁾) measured via the equation 23. If the data compensationprocedure using the first residual CFO is completed, the receptionmodule estimates a second residual CFO (Δ{circumflex over (ϵ)}_(n) ⁽²⁾)among the total residual CFO (Δ{circumflex over (ϵ)}_(n)) using theaforementioned first CFO estimation method or the second CFO estimationmethod. When data transmission is performed using QPSK, the procedure ofestimating the second residual CFO is represented as equation 26described in the following.

$\begin{matrix}{z_{k}^{n} = \left\{ \begin{matrix}y_{k}^{n} & {{{{if}\mspace{14mu} {{real}\left( y_{k}^{n} \right)}} \geq 0},{{{real}\left( y_{k}^{n} \right)} \geq {{imag}\left( y_{k}^{n} \right)}}} \\{- y_{k}^{n}} & {{{{if}\mspace{14mu} {{real}\left( y_{k}^{n} \right)}} < 0},{{{real}\left( y_{k}^{n} \right)} \geq {{imag}\left( y_{k}^{n} \right)}}} \\{{- j} \times y_{k}^{n}} & {{{{if}\mspace{14mu} {{imag}\left( y_{k}^{n} \right)}} \geq 0},{{{real}\left( y_{k}^{n} \right)} < {{imag}\left( y_{k}^{n} \right)}}} \\{j \times y_{k}^{n}} & {{{{if}\mspace{14mu} {{imag}\left( y_{k}^{n} \right)}} < 0},{{{real}\left( y_{k}^{n} \right)} < {{imag}\left( y_{k}^{n} \right)}}}\end{matrix} \right.} & \left\lbrack {{Equation}\mspace{14mu} 26} \right\rbrack\end{matrix}$

The equation 26 is identical to the equation 13 and indicates aprocedure for changing a phase of the total data into an in-phase stateto estimate the second residual CFO which has reduced to a small size.If BPSK or QPBSK is used instead of QPSK, the equation 6 is used for thesame procedure. Meanwhile, if BPSK and QBPSK are alternately used fortwo consecutive OFDM symbols, it may use the equation 11 instead of theequation 25. Or, if BPSK (or, QBPSK) and QPSK are alternately used fortwo consecutive OFDM symbols, it may use the equation 18 instead of theequation 25.

In addition to the equation 26, the second residual CFO is estimatedaccording to equation 27.

$\begin{matrix}{{\Delta \; {\hat{\epsilon}}_{n}^{(2)}} = {{angle}\left\{ {\sum\limits_{n = 1}^{L}{\sum\limits_{k \in C_{n}}z_{k}^{n}}} \right) \times \frac{N}{2\; {\pi \left( {N + N_{g}} \right)}}}} & \left\lbrack {{Equation}\mspace{14mu} 27} \right\rbrack\end{matrix}$

Unlike the equation 23, the equation 27 calculates a residual CFO forC_(n). In particular, a residual CFO is calculated for subcarriers atwhich data exists only except a subcarrier at which an RS exists. If thesecond residual CFO is finally calculated via the equation 27, the totalCFO can be represented by the sum of the first residual CFO and thesecond residual CFO (equation 28).

Δ{circumflex over (ϵ)}_(n)=Δ{circumflex over (ϵ)}_(n) ⁽¹⁾+Δ{circumflexover (ϵ)}_(n) ⁽²⁾  [Equation 28]

In the following, various embodiments capable of being applied to aprocedure for estimating a CFO are explained according to a series ofaforementioned procedures.

According to one embodiment, a reception module can calculate a residualCFO not only for a subcarrier at which data exists bit also for asubcarrier at which an RS exists in the course of performing a procedureof estimating a second residual CFO. In particular, the second residualCFO can be calculated via equation 29 rather than the equation 27.

$\begin{matrix}{{\Delta \; {\hat{\epsilon}}_{n}^{(2)}} = {{{angle}\left( {\sum\limits_{n = 1}^{L}\left( {{\sum\limits_{k \in C_{n}}z_{k}^{n}} + {\sum\limits_{k \in D_{n}}{y_{k}^{n}e^{{- j}\; 2\; \pi \; \Delta \; {{{\hat{\epsilon}}_{n}^{(1)}{({N + N_{g}})}}/N}}}}} \right)} \right)} \times \frac{N}{2\; {\pi \left( {N + N_{g}} \right)}}}} & \left\lbrack {{Equation}\mspace{14mu} 29} \right\rbrack\end{matrix}$

According to the abovementioned embodiment, since the number of samplesused for estimating the second residual CFO is increased as many as thenumber of subcarriers at which an RS exists, performance of estimatingthe second residual CFO is enhanced. For example, if the number of datasubcarriers is identical to the number of RS subcarriers (|C|=|D|),performance for estimating the second residual CFO according to theequation 29 is better than performance for estimating the secondresidual CFO according to the equation 27 as much as 3 dB. Moreover,since the second residual CFO is defined in a form of improving theaccuracy of the first residual CFO, performance for estimating the totalresidual CFO (or, a third residual CFO) is also enhanced as much as 3dB.

In the foregoing description, if the first residual CFO is calculated,the reception module compensates for data using the calculated firstresidual CFO and then calculates the second residual CFO. On thecontrary, according to one embodiment, a procedure of compensating fordata and a procedure of calculating the second residual CFO can beperformed by a single procedure in a manner of being integratedaccording to equation 30.

$\begin{matrix}{{\Delta \; {\hat{\epsilon}}_{n}} = {{{angle}\left( {\sum\limits_{n = 1}^{L}\left( {{\sum\limits_{k \in C_{n}}{z_{k}^{n}e^{j\; 2\; \pi \; \Delta \; {{{\hat{\epsilon}}_{n}^{(1)}{({N + N_{g}})}}/N}}}} + {\sum\limits_{k \in D_{n}}y_{k}^{n}}} \right)} \right)} \times \frac{N}{2\; {\pi \left( {N + N_{g}} \right)}}}} & \left\lbrack {{Equation}\mspace{14mu} 30} \right\rbrack\end{matrix}$

Specifically, if the first residual CFO is calculated, a third residualCFO can be immediately calculated without the procedure of compensatingfor data and the procedure of calculating the second residual CFO. Tothis end, a phase of a reception signal (z_(k) ^(n)) is changed as muchas the first residual CFO in the equation 30. And, according to thepresent embodiment, similar to the equation 29, the third residual CFOis calculated using both a data carrier and an RS carrier. The equation30 can be represented as equation 31 via mathematical calculation. Bydoing so, it is able to know that the total residual CFO calculated viathe equation 30 is identical to the total residual CFO according to theequations 22 to 28.

[Equation  31] $\begin{matrix}{{\Delta \; {\hat{\epsilon}}_{n}} = {{\Delta \; {\hat{\epsilon}}_{n}^{(1)}} + {{{angle}\left( {\sum\limits_{n = 1}^{L}\left( {{\sum\limits_{k \in C_{n}}z_{k}^{n}} + {\sum\limits_{k \in D_{n}}{y_{k}^{n}e^{{- j}\; 2\; \pi \; \Delta \; {{{\hat{\epsilon}}_{n}^{(1)}{({N + N_{g}})}}/N}}}}} \right)} \right)} \times}}} \\{\frac{N}{2\; \pi \; \left( {N + N_{g}} \right)}} \\{= {{\Delta \; {\hat{\epsilon}}_{n}^{(1)}} + {\Delta \; {\hat{\epsilon}}_{n}^{(2)}}}}\end{matrix}$

In the foregoing description, a CFO estimation method has been explainedunder a condition (D_(n)=D_(n+1)) that a subcarrier at which an RS ispositioned is identical to each other in two consecutive OFDM symbols.However, the condition may not be satisfied depending on a form ofdeploying data and an RS to a resource region. For example, it mayconsider such a case as D_(n)⊂D_(n+1) or D_(n+1)⊃D_(n).

In particular, when a reception module performs a procedure ofestimating the first residual CFO via an RS, the reception module canperform the procedure on a subcarrier only where an RS exists in both ann^(th) OFDM symbol and an (n+1)^(th) OFDM symbol. This is because, ifthe reception module knows either s_(k) ^(n) or s_(k) ^(n+1) only, aphase ambiguity problem occurs.

Moreover, if the first residual CFO is estimated according to acondition of D _(n)={D_(n)∩D_(n+1)}, C_(n) corresponding to a set ofsubcarriers at which data exists in an n^(th) OFDM symbol can also bedefined as C _(n)=C_(n)∪{D_(n)−{D_(n)∩D_(n+1)}}. In particular, when anRS exists in both two contiguous OFDM symbols in a subcarrier,subcarriers except the subcarrier are excluded from the procedure ofestimating the first residual CFO and the subcarriers are used assamples in the procedure of estimating the first residual CFO.

In the foregoing description, a CFO estimation method has been explainedon the basis of a subcarrier of which an RS is positioned at twoconsecutive OFDM symbols. Yet, if a subcarrier of which an RS ispositioned at two consecutive OFDM symbols does not exist, such a caseas D _(n)={ϕ} may occur. In this case, the aforementioned procedure forestimating the first residual CFO can be omitted. In particular, afollowing procedure can be performed by setting a first residual CFO(Δ{circumflex over (ϵ)}_(n) ⁽¹⁾) to 0.

On the contrary, the aforementioned first residual CFO estimationprocedure can also be applied to OFDM symbols away from each other asmuch as G. In particular, as shown in equation 32, it may be able tocalculate the first residual CFO for a subcarrier of which an RS existsin both an n^(th) OFDM symbol and an (n+G)^(th) OFDM symbol.

$\begin{matrix}{{{y_{k}^{n}\overset{\Delta}{=}{r_{k}^{n + G}\left( r_{k}^{n} \right)}^{*}},{k \in {\overset{\_}{D}}_{n,G}},{n = 1},2,\ldots \mspace{14mu},L}{{\Delta \; {\hat{\epsilon}}_{n}^{(1)}} = {{{angle}\left( {\sum\limits_{n = 1}^{L}{\sum\limits_{k \in D_{n}}{y_{k}^{n}\left( s_{k}^{n + G} \right)}^{*}}} \right)} \times \frac{N}{2\; \pi \; {G\left( {N + N_{g}} \right)}}}}} & \left\lbrack {{Equation}\mspace{14mu} 32} \right\rbrack\end{matrix}$

In the equation 32, D _(n,G)≙{D_(n)∩D_(n+G)} is defined. In other word,in the foregoing description, a CFO estimation method has been explainedunder a condition that a channel and a CFO do not change in twocontiguous OFDM symbols. On the contrary, referring to the equation 32,a reception module estimates a CFO under a condition that a channel anda CFO do not change during G+1 number of OFDM symbols. An error e_(k)^(n) according to the abovementioned procedure can be defined asequation 33 in the following.

e _(k) ^(n) ≙|y _(k) ^(n) −|H _(k) ^(n)|² s _(k) ^(n+G)(s _(k) ^(n))*e^(j2πΔϵ) ^(n) ^((N+N) ^(g) ^()G/N)|²  [Equation 33]

The error e_(k) ^(n) increases as a distance (G) between two OFDMsymbols is getting bigger. Hence, in the present embodiment, if D_(n,G)≙{D_(n)∩D_(n+G)}≠{ϕ} is satisfied and a channel and a CFO do notconsiderably change during G+1 number of OFDM symbols, efficiency ofestimating a CFO can be increased.

In the foregoing description, a CFO is estimated on a subcarrier commonto two OFDM symbols (two contiguous OFDM symbols or OFDM symbols awayfrom each other as much as G). Yet, according to a different embodiment,a CFO can also be estimated via contiguous subcarriers rather than thesame subcarrier. For example, a procedure for estimating the firstresidual CFO can be changed like equation 34 described in the following.

y _(k) ^(n) ≙r _(k+δ) _(D) ^(n+1)(r _(k) ^(n))*, k∈D _(n) , n=1,2, . . .,L  [Equation 34]

In the equation 34, δ_(D) corresponds to a distance between twosubcarriers used for estimating a CFO. If the first residual CFO iscalculated based on the equation 34, a procedure for calculating thesecond residual CFO can be changed like equation 35 described in thefollowing.

y _(k) ^(n) ≙r _(k+δ) _(C) ^(n+1)(r _(k) ^(n))*e^(−j2πΔ{tilde over (ϵ)}) ^(n) ⁽¹⁾ ^((N+N) ^(g) ^()/N) , k∈D _(n) ,n=1,2, . . . ,L  [Equation 35]

In the equation 35, δ_(C) also corresponds to a distance betweenadjacent subcarriers. In the equations 34 and 35, assume that channelsbetween adjacent subcarriers are identical to each other (H_(k+δ)^(n+1)=H_(k) ^(n)). An error caused by the assumption is represented byequation 36 in a manner of being defined by an equation similar to theequation 33.

e _(k) ^(n) ≙|y _(k) ^(n) −|H _(k) ^(n)|² s _(k) ^(n+1)(s _(k) ^(n))*e^(−j2πΔϵ) ^(n) ^((N+N) ^(g) ^()/N)|²  [Equation 36]

where H_(k+δ) ^(n+1)(H_(k) ^(n))*≈|H_(k) ^(n)|²

An error e_(k) ^(n) increases as a difference between H_(k) ^(n) andH_(k+δ) ^(n+1) is getting bigger. Hence, in the present embodiment, asselectivity of a frequency axis is getting smaller, efficiency isincreasing. For example, if a delay profile of a channel is smallsimilar to in-door environment, the abovementioned embodimentefficiently works.

Meanwhile, in the equations 34 and 35, the δ_(C) and the δ_(D) can beconfigured by a different value. If the two values are configured by 0,the equations 34 and 35 are identical to the equations 22 and 25.

According to the third proposed method for CFO estimation, a receptionmodule estimates a first residual CFO using an RS known to the receptionmodule in advance. The first residual CFO is used for a procedure ofcompensating for data. A second residual CFO is calculated from thecompensated data according to the first CFO estimation method or thesecond CFO estimation method.

Meanwhile, the procedure for estimating the first residual CFO using anRS has been explained under the condition that a channel is common totwo OFDM symbols. Yet, in case of Case A shown in FIG. 28, CRSs aretransmitted via a different antenna port in a first OFDM symbol and asecond OFDM symbol. In particular, since channels are different fromeach other in the two OFDM symbols, it is unable to apply theabovementioned scheme as it is. Yet, in case of E-PDCCH (evolvedphysical downlink control channel) in LTE/LTE-A, since two OFDM symbolsare transmitted via the same antenna port, it may be able to apply theproposed CFO estimation method as it is.

Referring to FIG. 28, an RS is not assigned to a 3^(rd) OFDM symbol anda 4^(th) OFDM symbol. Hence, the total residual CFO is estimated withouta procedure for estimating the first residual CFO. Among the 4^(th) OFDMsymbol and a 5^(th) OFDM symbol, since an RS exists at the 5^(th) OFDMsymbol only, D _(n)={ϕ} is satisfied. Hence, the procedure forestimating the first residual CFO is omitted.

Meanwhile, in case of Case B shown in FIG. 28 (6^(th) and 7^(th) OFDMsymbols), RSs twice as many as RSs of the Case A are assigned. Inparticular, since the Case B has double samples for estimating the firstresidual CFO, performance of estimating the first residual CFO isimproved as much as 3 dB compared to the Case A.

In case of Case C (9^(th) and 11^(th) OFDM symbols), it may be able toestimate the first residual CFO according to the equation 33 on thebasis of OFDM symbols away from each other as much as G=1. Yet, it ishighly probable that estimation performance is to be deterioratedcompared to the Cases A and B.

According to the aforementioned embodiments, a reception moduleestimates the first residual CFO using data known to the receptionmodule in advance to compensate for a part of CFOs of the data (a phaseambiguity problem does not occur in this procedure) and estimates thesecond residual CFO according to the aforementioned blind type CFOestimation method. Since it is able to apply the blind type CFOestimation method to a case that a size of a residual CFO is small,performance and efficiency in estimating a residual CFO can be enhanced.

FIG. 29 is a flowchart for a method of estimating a CFO according to thepresent invention.

FIG. 29 illustrates the CFO estimation method according to theembodiments mentioned earlier in FIG. 28 according to a time serialflow. Hence, although the embodiments mentioned earlier in FIG. 28 arenot explained in detail in FIG. 29, the embodiments can be similarly oridentically applied.

First of all, a transmission module transmits data to a reception module[S2910]. The data can be transmitted in a unit of a frame defined by anOFDM symbol and a subcarrier. The data is transmitted to the receptionmodule in a manner of being mapped to a specific constellation. Theconstellation may use BPSK, QBPSK, QPSK, etc. BPSK and QBPSK can bealternately used for a series of contiguous (or consecutive) OFDMsymbols or BPSK (or, QBPSK) and QPSK can be alternately used for thecontiguous OFDM symbols.

Meanwhile, the reception module firstly estimates a CFO value from areception signal [S2920]. This procedure is performed using such apreamble part as L-STF, L-LTF, and the like of a frame. Yet, since theCFO value changes over time, the CFO value estimated in the step S2920may not be an accurate CFO value. Hence, the reception module performs aprocedure for estimating a residual CFO in succession.

The reception module estimates a first residual CFO value using an RS[S2930]. A procedure for estimating the first residual CFO value isperformed by measuring the first residual CFO using RSs received fromtwo contiguous OFDM symbols on a specific subcarrier. The receptionmodule compensates for a phase value of data using the first residualCFO and estimates a second residual CFO on the basis of the compensateddata [S2940]. Since the data is compensated via the first residual CFOestimated in the step S2930, the second residual CFO estimated in thestep S2940 may have a relatively small value. In particular, thereception module estimates the second residual CFO according to the stepS2330 of FIG. 23 and the step S2730 of FIG. 27. The sum of the tworesidual CFOs estimated in the steps S2930 and S2940 becomes the totalresidual CFO.

Lastly, the reception module is able to precisely decode the datatransmitted by the transmission module by eliminating the impact of theCFO estimated in the step S2920 and the total residual CFO estimated inthe steps S2930 and S2940.

6. Fourth Proposed Method for CFO Estimation

In the foregoing description, embodiments for a reception module tocalculate a residual CFO using the first, the second and the third CFOestimation method have been explained. Yet, since the embodimentsexplain a case of transmitting data using BPSK, QBPSK, and QPSKmodulation scheme only, it is unable to apply the embodiments to 16-QAM(16-Quadrature Amplitude modulation) corresponding to a modulationscheme of a higher order, and the like. This is because theaforementioned blind type CFO estimation method is performed under acondition that a phase difference between signals received from twocontiguous OFDM symbols on a specific subcarrier is gathered to aspecific value. If a modulation scheme of an order equal to or higherthan an order of the 16-QAM modulation scheme is used, the condition isnot valid anymore.

Hence, when data is transmitted using a modulation scheme of a highorder (e.g., 16-QAM), a method for a reception module to estimate a CFOis explained in the following with reference to FIGS. 30 to 32.

First of all, as mentioned earlier in the proposed embodiments, a pairof received signals corresponds to signals received from two contiguousOFDM symbols. In the equation 5, C indicates a set of subcarrier indexesof the total reception signal pairs and the aforementioned CFOestimation performance is proportional to a size of the set C. Inparticular, as the number of samples (reception signal pairs) applied toa CFO estimation algorithm is getting bigger, the estimation performanceis more enhanced. As mentioned in the foregoing description, if datatransmission is performed using BPSK, QBPSK, and QPSK, all receptionsignal pairs are used for estimating a CFO.

Meanwhile, if the CFO estimation method is applied to a high order QAM(e.g., 16-QAM), only a part of the total reception signal pairs can beused as a sample. Hence, the equation 5 can be represented as equation37 descried in the following.

y _(k) ^(n) ≙r _(k) ^(n+1)(r _(k) ^(n))*, k∈C, n=1,2, . . . ,L[Equation37]

where C⊂C

In the equation 37, C corresponds to a subset selected from the Ccorresponding to the set of total reception signal pairs via anembodiment described in the following. In particular, an object of theproposed embodiment is to maximize performance by increasing the numberof samples for estimating a CFO through a method of selecting areception signal pair for maximizing a size of the C.

FIGS. 30 and 31 are diagrams for a method of segmenting 16-QAMconstellation according to embodiments proposed in the presentinvention. As shown in FIGS. 30 and 31, a part of points located on16-QAM constellation is matched with points located on QPSKconstellation. Hence, if signals received from two OFDM symbols aretransmitted in a manner of being mapped to points corresponding to QPSKamong the points located on the 16-QAM constellation, it may be able toapply the first, the second, and the third CFO estimation methods as itis. Based on this, a method of estimating a CFO using a size and a phaseof a reception signal pair is explained in the following.

First of all, according to the proposed embodiment, a reception modulecompares sizes between a pair of received signals and determines whetherto use the pair of received signals in a procedure for estimating aresidual CFO. When the reception module compares the sizes between thepair of received signals, the reception module can calculate at leastone of a ratio of the sizes and a difference between the sizes. Equation38 illustrates a procedure of comparing the ratio of the sizes of thepair of received signals and equation 39 illustrates a procedure ofcomparing the difference between the sizes of the pair of receivedsignals.

$\begin{matrix}{\gamma_{k}^{n}\overset{\Delta}{=}{\frac{r_{k}^{n + 1}}{r_{k}^{n}}\mspace{14mu} {or}\mspace{14mu} {\frac{r_{k}^{n + 1}}{r_{k}^{n}}}}} & \left\lbrack {{Equation}\mspace{14mu} 38} \right\rbrack \\{\kappa_{k}^{n}\overset{\Delta}{=}{{r_{k}^{n + 1}} - {{r_{k}^{n}}\mspace{14mu} {or}\mspace{14mu} {{r_{k}^{n + 1} - r_{k}^{n}}}}}} & \left\lbrack {{Equation}\mspace{14mu} 39} \right\rbrack\end{matrix}$

In equations 38 and 39, r_(k) ^(n) corresponds to a signal received froman n^(th) OFDM symbol and k indicates subcarrier indexes at which then^(th) OFDM symbol and an (n+1)^(th) OFDM symbol are located. In theequations 38 and 39, (r_(k) ^(n), r_(k) ^(n+1)) corresponds to theaforementioned pair of received signals.

Meanwhile, in consideration of a condition that two channels on the samesubcarrier are identical to each other in the n^(th) OFDM symbol and the(n+1)^(th) OFDM symbol and a CFO and noise do not exist in the n^(th)OFDM symbol and the (n+1)^(th) OFDM symbol, the equation 38 can berepresented as equation 40 described in the following.

$\begin{matrix}{\gamma_{k}^{n}\overset{\Delta}{=}{\frac{r_{k}^{n + 1}}{r_{k}^{n}} = {\frac{{H_{k}^{n}s_{k}^{n + 1}}}{{H_{k}^{n}s_{k}^{n}}} = {\frac{{H_{k}^{n}}\; {s_{k}^{n + 1}}}{{H_{k}^{n}}\; {s_{k}^{n}}} = \frac{s_{k}^{n + 1}}{s_{k}^{n}}}}}} & \left\lbrack {{Equation}\mspace{14mu} 40} \right\rbrack\end{matrix}$

In particular, since a channel is the same in two OFDM symbols, a sizeratio of a pair of received signals received by a reception module canbe represented by a size ratio of data symbols s_(k) ^(n),s_(k) ^(n+1)transmitted by a transmission module. If sizes of data symbolstransmitted by the transmission module are the same, a size ratio valueof a pair of received signals becomes 1.

FIG. 30 illustrates an embodiment of segmenting 16-QAM constellationinto 3. In particular, if (A), (b), and (C) shown in FIG. 30 are summedup, it becomes 16-QAM constellation. In particular, 16-QAM constellationis segmented into (A), (b), and (C) according to a data symbol havingthe same power. Hence, if a size ratio γ_(k) ^(n) of a pair of signalsreceived from a reception module corresponds to 1, it is able to knowthat two data symbols constructing the pair of the received signalsbelong to at least one of (A), (b), and (c).

For example, if the size ratio of the pair of the received signalscorresponds to 1, the two data symbols may correspond to one selectedfrom the group consisting of (a1, a2), (a1, a4), (b1, b2), and (c1, c3).However, the two data symbols are unable to become (a1, b1) or (b1, c1).Consequently, a reception module is able to find out a pair of receivedsignals corresponding to two data symbols having the same power using asize ratio of the pair of the received signals without using channelinformation (i.e., using a non-coherent scheme).

Meanwhile, in FIG. 30, (b) and (c) have an arrangement form identical toQPSK constellation, respectively. Hence, if it is determined that thepair of the received signals exists in one of (b) and (c) at the sametime, the reception module is able to estimate a residual CFO using theaforementioned second CFO estimation method.

On the contrary, the pair of the received signals may belong to (A). Aphase difference between data symbols can be classified into two typesin the (A) group. First of all, a phase difference between pointsexpressed by the same marker (quadrangle or circle) corresponds to oneselected from the group consisting of {0, π/2, π, 3π/2}. And, a phasedifference between points expressed by a different marker corresponds toone selected from the group consisting of {46°, 136°, 226°, 316°}.

In case of the first case (points expressed by the same marker), similarto (B) and (C), since a phase difference between symbols is identical tothat of QPSK, it may be able to similarly apply the second CFOestimation method (hereinafter, (A-1) case). On the contrary, in case ofthe second case (points expressed by a different marker), since a phasedifference is different between received signals, it is unable to applythe aforementioned CFO estimation methods as it is (hereinafter, (A-2)case). In case of the (A-1), the (b), and the (C), a reception module isable to immediately estimate a residual CFO through the second CFOestimation method. However, in case of the (A-2) case, if the receptionmodule changes a phase of received signals as much as −46°, thereception module can process the (A-2) case in a manner of beingidentical to the case of the (A-1) case. In particular, it is necessaryfor the reception module to distinguish the (A-1) case from the (A-2)case. Regarding this, it shall be described later.

FIG. 30 illustrates a different embodiment of segmenting 16-QAMconstellation into 2. Similar to FIG. 30, if (A) and (B) shown in FIG.31 are summed up, it becomes 16-QAM constellation. If a size ratio of apair of received signals corresponds to ⅓ or 3, a size differencebetween two data symbols selected by a transmission module has adifference as much as three times. In this case, the data symbols of thepair of the received signals belong to FIG. 31 (b) at the same time.FIG. 31 (b) corresponds to the sum of FIGS. 30 (b) and (c). Hence,although a size ratio of a pair of received signals corresponds to ⅓ or3, a reception module is able to estimate a CFO by applying the secondCFO estimation method. In other word, when a size ratio of a pair ofreceived signals corresponds to ⅓ or 3, if the reception module appliesthe second CFO estimation method, the number of samples used forestimating a CFO increases (i.e., a size of C increases) compared to acase of applying the second CFO estimation method only when a size ratiocorresponds to 1, thereby enhancing CFO estimation performance. Forexample, although a size ratio of a pair of received signals correspondsto ⅓ or 3, if the reception module allows the ratio to be used as asample for a method of estimating a CFO, the reception module is able toestimate a CFO using such a pair of received signals as (b1, c2) and(b2, c2) shown in FIG. 31 (B). (Yet, such a pair of received signals as(a1, b1) cannot be used as a CFO estimation candidate.)

The aforementioned embodiment is summarized in the following. If a sizeratio of a pair of received signals belongs to a prescribed range, areception module uses the pair of received signals in the course ofperforming a procedure for estimating a residual CFO. According to oneembodiment, a pair of received signals can be selected on the basis of asize difference rather than the size ratio of the pair of receivedsignals.

In particular, theoretically, although it is accurate to determinewhether or not a size ratio of a pair of received signals is identicalto a specific value, when a system is actually implemented, channels onwhich the pair of received signals is transmitted are not the perfectlysame. And, an impact of noise and a CFO exist all the time. Hence, it isdifficult to apply the propose embodiment only when the size ratio ofthe pair of received signals is perfectly matched with a prescribedvalue. Hence, if it is determined that the size ratio has a valuebelonging to a prescribed range of the prescribed value, it may applythe proposed embodiment. In particular, if a size ratio of a pair ofreceived signals satisfies a range of equation 41, the pair of receivedsignals can be used in the course of performing a procedure forestimating a CFO.

1−δ≤γ_(k) ^(n)≤1+δ or 3−δ≤γ_(k) ^(n)≤3+δ  [Equation 41]

In the equation 41, δ corresponds to a threshold range for which a sizeratio of a pair of received signals is allowed. For example, the δ canbe configured by 0.1.

In the following, an embodiment for distinguishing the (A-1) case fromthe (A-2) case is additionally explained. As mentioned earlier in FIG.30, if a size ratio of a pair of received signals corresponds to 1, (b)and (C) can be utilized as samples for performing the second CFOestimation method. Not only the (A-1) case but also the (A-2) caserequires a procedure of changing phases of received signals. Hence, amethod of identifying the (A-2) case is explained in the following.

According to an embodiment proposed in the present invention, areception module can determine whether to use a pair of received signalsas a sample for performing a method of estimating a residual CFO inconsideration of a phase difference of the pair of received signals. Amethod of calculating a phase of a pair of received signals received bya reception module using a phase of a data symbol transmitted by atransmission module is explained before the embodiment is explained. Inthe aforementioned second CFO estimation method, a phase of y_(k) ^(n)corresponds to a phase of multiplication of a pair of signals receivedby a reception module. In this case, the phase is identical to a phaseof multiplication of data symbols transmitted by a transmission module.This is because, when the pair of received signals is multiplied, phasesof channels are cancelled out. Hence, it is able to know that a phase ofa channel experienced by a reception signal does not influence on thephase of multiplication of the pair of received signals. In particular,although the phase of multiplication of the pair of received signals isreplaced with the phase of multiplication of the data symbolstransmitted by the transmission module irrespective of a channel of areceived signal, there is no problem.

Meanwhile, in order to identify the (A-2) case of FIG. 30, the receptionmodule preferentially checks whether or not the pair of received signalssatisfies the equation 41. If the pair of received signals satisfies theequation 41, the reception module checks whether or not a phasedifference of the pair of received signals satisfies equation 42described in the following.

α−δ≤λ_(k) ^(n)≤α+δ  [Equation 42]

where λ_(k) ^(n)=

(r_(k) ^(n+1)(r_(k) ^(n))*)

In the equation 42, α∈{0°, 90°, 180°, 270° } is satisfied and δcorresponds to a prescribed threshold range. For example, if the αcorresponds to 90° and the δ corresponds to 10°, the equation 42corresponds to 80°≤λ_(k) ^(n)≤100°. If the equation 42 is satisfied, areception module is able to know that the pair of received signalscorresponds to the (A-1), (B), and (C) cases and the pair of receivedsignals can be used as a sample for estimating a residual CFO.Meanwhile, if the equation 42 is not satisfied, the reception module isable to know that the pair of received signals corresponds to the (A-2)case. In this case, the reception module does not use the pair ofreceived signals for the CFO estimation procedure. Meanwhile, candidatevalues for the α are not restricted to 0°, 90°, 180°, and 270°. Thecandidate values can be configured by values of a different rangeadjacent to 0°, 90°, 180°, and 270°.

In case of the (A-2) case of FIG. 30, as mentioned in the foregoingdescription, a phase difference of the pair of received signals maycorrespond to one selected from the group consisting of {46°, 136°,226°, 316°}. Hence, if the phase difference of the pair of receivedsignals is calculated as being adjacent to a specific value, the pair ofreceived signals can be used for estimating a CFO via a simpleadditional procedure.

According to an embodiment proposed in the present invention, if α∈{0°,90°, 180°, 270°} and α∈{46°, 136°, 226°, 316°} are determined in theequation 42 and a phase difference λ_(k) ^(n) is determined as α∈{46°,136°, 226°, 316°}, the reception module may change a phase ofmultiplication of the pair of received signals as much as

$e^{{- j}\frac{46{^\circ}}{180{^\circ}}\pi}$

according to equation 43 described in the following. Meanwhile, 46°,136°, 226°, and 316° are just an example only. It may use differentvalues adjacent to the 46°, 136°, 226°, and 316°.

$\begin{matrix}{y_{k}^{n} = {{r_{k}^{n + 1}\left( r_{k}^{n} \right)}^{*}e^{{- j}\frac{46{^\circ}}{180{^\circ}}\pi}}} & \left\lbrack {{Equation}\mspace{14mu} 43} \right\rbrack\end{matrix}$

According to the equation 43, a changed phase difference of the pair ofreceived signals may correspond to one selected from the groupconsisting of 0°, 90°, 180°, and 270°. Hence, the phase differencechanged by the equation 43 becomes similar to the (A-1) case and thepair of received signals can be used for estimating a CFO. Inparticular, if the phase of multiplication of the pair of receivedsignals is changed, the number of pairs of received signals capable ofbeing used for estimating a CFO increase. In particular, since a size ofC increases, it may be able to enhance CFO estimation performance.

In the aforementioned embodiments, a reception module calculates a sizeratio (or, a size difference) and a phase difference of a pair ofreceived signals to determine whether or not the pair of received signalis used for estimating a CFO. Table 2 in the following illustrates that|C| varies according to a method for a reception module to manage anembodiment proposed in the present invention. Table 2 illustratesprobability values that vary according to each of methods.

TABLE 2     Size ratio of pair of received signals Whether or not phasedifference of pair of received signals is compensated (whether or notnot phase changing embodiment is applied)  $p\left( \frac{\overset{\_}{C}}{C} \right)$ γ_(k = 1) ^(n) X 1/4γ_(k = 1) ^(n) O 3/8 γ_(k = 1 or 3) ^(n) X 3/8 γ_(k = 1 or 3) ^(n) O 1/2

In Table 2, if γ_(k) ^(n) corresponds to 1 and a phase changingembodiment is not applied, (A-1), (b), and (C) of FIG. 30 are allowedonly as samples for performing a CFO estimation method. In this case, aprobability that two data symbols corresponding to a pair of receivedsignals belong to (b) or (C) at the same time corresponds to (¼)*(¼)=(1/16). A probability that two data symbols corresponding to a pair ofreceived signals belong to (a) only corresponds to (½)*(½)=(¼). Since(A-1) is processed only, a probability for the (A-1) becomes (⅛).Consequently, a value of

$p\mspace{11mu} \left( \frac{\overset{\_}{C}}{C} \right)$

indicating a ratio of a pair of received signals, which are used forestimating a CFO, among the total reception signal pairs becomes (1/16)+( 1/16)+(⅛)=(¼).

Similarly, if γ_(k) ^(n) corresponds to 1 and a phase changingembodiment is applied, (A-1), (A-2), (b), and (C) are all allowed assamples for performing a CFO estimation method. In this case, a value of

$p\mspace{11mu} \left( \frac{\overset{\_}{C}}{C} \right)$

becomes ( 1/16)+( 1/16)+(¼)=(⅜).

If γ_(k) ^(n) corresponds to 1 or 3 and a phase changing embodiment isnot applied, (A-1) of FIG. 30 and (b) of FIG. 31 are allowed as samplesfor performing a CFO estimation method. In this case, a probability thattwo data symbols corresponding to a pair of received signals belong to(b) of FIG. 31 at the same time corresponds to (½)*(½)=(¼). Aprobability that two data symbols corresponding to a pair of receivedsignals belong to (A-1) corresponds to (⅛). Consequently, a value of

$p\mspace{11mu} \left( \frac{\overset{\_}{C}}{C} \right)$

becomes (¼)+(⅛)=(⅜).

If γ_(k) ^(n) corresponds to 1 or 3 and a phase changing embodiment isapplied, (A-1) and (A-2) of FIG. 30 and (b) of FIG. 31 are allowed assamples for performing a CFO estimation method. Consequently, a value of

$p\mspace{11mu} \left( \frac{\overset{\_}{C}}{C} \right)$

becomes (¼)+(¼)=(½).

In particular, a reception module calculates a size ratio and a phasedifference value of a pair of received signals and compares thecalculated value with a predetermined number to determine whether or notthe pair of received signals is used for a method of estimating a CFO.If more pairs of received signals are used for a CFO estimation method,CFO estimation performance can be increased. As mentioned in theforegoing description, the reception module may utilize the half of thetotal reception signal pairs according to 16-QAM for CFO estimation.Meanwhile, information on a maximum value for which a size ratio isallowed and information on whether or not a phase difference iscompensated can be determined by a user in advance or can besystematically determined. For example, if channel environment is notsufficiently good, a reception module may allow a size ratio of 1 onlyto make a phase difference compensation not to be applied. Inparticular, the number of pairs of received signals utilized for a CFOestimation method can be adaptively changed according to communicationenvironment.

In the foregoing description, an embodiment of similarly applying ablind type residual CFO estimation method, which was applied to BPSK,QBPSK, and QPSK, has been explained by segmenting 16-QAM. In this case,16-QAM is just an example only for clarity. The proposed method can alsobe similarly applied to various modulation schemes such as 32, 64, 128,and 256 QAM.

FIG. 32 is a flowchart for a method of estimating a CFO according to thepresent invention. FIG. 32 illustrates the CFO estimation methodaccording to the embodiments mentioned earlier in FIGS. 30 and 31according to a time serial flow. Hence, although the embodimentsmentioned earlier in FIGS. 30 and 31 are not explained in detail in FIG.32, the embodiments can be similarly or identically applied.

First of all, a transmission module transmits data to a reception module[S3210]. The data can be transmitted in a unit of a frame defined by anOFDM symbol and a subcarrier. The data is transmitted to the receptionmodule in a manner of being mapped to a specific constellation. Theconstellation may use various high order QAMs including 16, 32, 64, 128,and 256 QAM. The present embodiment is explained using an example of16-QAM.

The reception module firstly estimates a CFO value from a receivedsignal [S3220]. This procedure is performed using such a preamble partas L-STF, L-LTF, and the like of a frame. Yet, since the CFO valuechanges over time, the CFO value estimated in the step S3220 may not bean accurate CFO value. Hence, the reception module performs a procedurefor estimating a residual CFO in succession.

First of all, the reception module selects a pair of received signals tobe used for estimating a residual CFO value [S3230]. This procedure canbe performed via a procedure for the reception module to calculate asize ratio of the pair of received signals. In addition to the procedureof calculating the size ratio of the pair of received signals, thereception module can also perform a procedure of calculating a phasedifference of the pair of received signals. If the calculated size ratioand/or the phase difference corresponds to a specific value or a valuebelonging to a threshold range adjacent to the specific value, thereception module determines to use the pair of received signals forestimating a residual CFO. In particular, pairs of received signals areselected in the step S3230 as samples for estimating a residual CFO.

Subsequently, the reception module estimates a residual CF using theselected pairs of received signals [S3240]. This procedure can beperformed via the second CFO estimation method mentioned earlier inFIGS. 24 to 26. In particular, the reception module interprets a part ofa pair of received signals received through 16-QAM as QPSK and estimatesa residual CFO using a residual CFO estimation method applied to a QPSKmodulation scheme.

Lastly, the reception module eliminates a CFO estimated from a receivedsignal and an impact of a residual CFO value to correctly decode thedata transmitted by the transmission module.

7. Apparatus Configuration

FIG. 33 is a block diagram showing the configuration of a receptionmodule and a transmission module according to one embodiment of thepresent invention. In FIG. 33, the reception module 100 and thetransmission module 200 may include radio frequency (RF) units 110 and210, processors 120 and 220 and memories 130 and 230, respectively.Although a 1:1 communication environment between the reception module100 and the transmission module 200 is shown in FIG. 33, a communicationenvironment may be established between a plurality of reception moduleand the transmission module. In addition, the transmission module 200shown in FIG. 33 is applicable to a macro cell base station and a smallcell base station.

The RF units 110 and 210 may include transmitters 112 and 212 andreceivers 114 and 214, respectively. The transmitter 112 and thereceiver 114 of the reception module 100 are configured to transmit andreceive signals to and from the transmission module 200 and otherreception modules and the processor 120 is functionally connected to thetransmitter 112 and the receiver 114 to control a process of, at thetransmitter 112 and the receiver 114, transmitting and receiving signalsto and from other apparatuses. The processor 120 processes a signal tobe transmitted, sends the processed signal to the transmitter 112 andprocesses a signal received by the receiver 114.

If necessary, the processor 120 may store information included in anexchanged message in the memory 130. By this structure, the receptionmodule 100 may perform the methods of the various embodiments of thepresent invention.

The transmitter 212 and the receiver 214 of the transmission module 200are configured to transmit and receive signals to and from anothertransmission module and reception modules and the processor 220 arefunctionally connected to the transmitter 212 and the receiver 214 tocontrol a process of, at the transmitter 212 and the receiver 214,transmitting and receiving signals to and from other apparatuses. Theprocessor 220 processes a signal to be transmitted, sends the processedsignal to the transmitter 212 and processes a signal received by thereceiver 214. If necessary, the processor 220 may store informationincluded in an exchanged message in the memory 230. By this structure,the transmission module 200 may perform the methods of the variousembodiments of the present invention.

The processors 120 and 220 of the reception module 100 and thetransmission module 200 instruct (for example, control, adjust, ormanage) the operations of the reception module 100 and the transmissionmodule 200, respectively. The processors 120 and 220 may be connected tothe memories 130 and 230 for storing program code and data,respectively. The memories 130 and 230 are respectively connected to theprocessors 120 and 220 so as to store operating systems, applicationsand general files.

The processors 120 and 220 of the present invention may be calledcontrollers, microcontrollers, microprocessors, microcomputers, etc. Theprocessors 120 and 220 may be implemented by hardware, firmware,software, or a combination thereof.

If the embodiments of the present invention are implemented by hardware,Application Specific Integrated Circuits (ASICs), Digital SignalProcessors (DSPs), Digital Signal Processing Devices (DSPDs),Programmable Logic Devices (PLDs), Field Programmable Gate Arrays(FPGAs), etc. may be included in the processors 120 and 220.

Meanwhile, the aforementioned method may be implemented as programsexecutable in computers and executed in general computers that operatethe programs using computer readable media. In addition, data used inthe aforementioned method may be recorded in computer readable recordingmedia through various means. It should be understood that programstorage devices that can be used to describe storage devices includingcomputer code executable to perform various methods of the presentinvention do not include temporary objects such as carrier waves orsignals. The computer readable media include storage media such asmagnetic recording media (e.g. ROM, floppy disk and hard disk) andoptical reading media (e.g. CD-ROM and DVD).

It will be apparent to those skilled in the art that variousmodifications and variations can be made in the present inventionwithout departing from the spirit or scope of the inventions. Thus, itis intended that the present invention covers the modifications andvariations of this invention provided they come within the scope of theappended claims and their equivalents.

INDUSTRIAL APPLICABILITY

Although the CFO estimation method is mainly described based on an IEEE802.11 system and an HEW system, the method can be applied to variouskinds of wireless communication systems as well as the IEEE 802.11system and the HEW system.

What is claimed is:
 1. A method of estimating a CFO (carrier frequency offset), which is estimated by a reception module in a wireless communication system using 16-QAM (quadrature amplitude modulation), the method comprising: measuring a size ratio between a pair of received signals received on two consecutive OFDM (orthogonal frequency division multiplexing) symbols for a specific subcarrier; when the size ratio belongs to a first threshold range, determining the pair of received signals as a pair to be used for estimating a residual CFO; and estimating the residual CFO using only pairs of received signals which are determined to be used for estimating the residual CFO.
 2. The method of claim 1, further comprising: measuring a phase difference between the pair of received signals, wherein when the size ratio belongs to the first threshold range and the phase difference belongs to a second threshold range, the pair of received signals is determined as a pair to be used for estimating the residual CFO.
 3. The method of claim 2, further comprising: when the phase difference does not belong to the second threshold range and belongs to a third threshold range, changing a phase of multiplication of the pair of received signals as much as a predetermined value.
 4. The method of claim 3, further comprising: when the phase difference does not belong to the second threshold range and does not belong to the third threshold range, determining not to use the pair of received signals for estimating the residual CFO.
 5. The method of claim 3, wherein the second threshold range indicates a range within a predetermined value from one value among 0°, 90°, 180°, and 270°.
 6. The method of claim 3, wherein the third threshold range indicates a range within a predetermined value from one value among 46°, 136°, 226°, and 316°.
 7. The method of claim 1, wherein the first threshold range indicates a range within a predetermined value from one value among 1, ⅓, and
 3. 8. The method of claim 1, wherein the step of estimating the residual CFO estimates the residual CFO according to a blind scheme which is applied when QPSK (quadrature phase shift keying) is used.
 9. A reception module estimating a CFO (carrier frequency offset) in a wireless communication system using 16-QAM (quadrature amplitude modulation), the reception module comprising: a transmitter; a receiver; and a processor operates in a manner of being connected with the transmitter and the receiver, wherein the processor: measures a size ratio between a pair of received signals received on two consecutive OFDM (orthogonal frequency division multiplexing) symbols for a specific subcarrier, the processor, when the size ratio belongs to a first threshold range, determines the pair of received signals as a pair to be used for estimating a residual CFO, and estimates the residual CFO using only pairs of received signals which are determined to be used for estimating the residual CFO.
 10. The reception module of claim 9, wherein the processor measures a phase difference between the pair of received signals, and wherein when the size ratio belongs to the first threshold range and the phase difference belongs to a second threshold range, the processor determines the pair of received signals as a pair to be used for estimating the residual CFO.
 11. The reception module of claim 10, wherein when the phase difference does not belong to the second threshold range and belongs to a third threshold range, the processor changes a phase of multiplication of the pair of received signals as much as a predetermined value.
 12. The reception module of claim 11, wherein when the phase difference does not belong to the second threshold range and does not belong to the third threshold range, the processor determines not to use the pair of received signals for estimating the residual CFO.
 13. The reception module of claim 11, wherein the second threshold range indicates a range within a predetermined value from one value among 0°, 90°, 180°, and 270°.
 14. The reception module of claim 11, wherein the third threshold range indicates a range within a predetermined value from one value among 46°, 136°, 226°, and 316°.
 15. The reception module of claim 9, wherein the first threshold range indicates a range within a predetermined value from one value among 1, ⅓, and
 3. 16. The reception module of claim 9, wherein the processor is configured to estimate the residual CFO according to a blind scheme which is applied when QPSK (quadrature phase shift keying) is used. 